Apparatus for transmitting and receiving a signal and method of transmitting and receiving a signal

ABSTRACT

A method for transmitting and receiving a signal and an apparatus for transmitting and receiving a signal are disclosed. The method for receiving the signal includes receiving (S 210 ) the signal in a first frequency band, identifying (S 220 ) a first pilot signal including, a cyclic prefix obtained by frequency-shifting a first portion of an useful portion of the first pilot signal and a cyclic suffix obtained by frequency-shifting a second portion of the useful portion of the first pilot signal from the received signal, demodulating (S 220 ) a signal frame including a physical layer pipe (PLP) to which a service stream is converted, by an orthogonal frequency division multiplexing (OFDM) scheme, using information set in the first pilot signal, parsing (S 230 ) the signal frame and obtaining the PLP and obtaining (S 240 ) the service stream from the PLP.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No. 13/620,612, filed on Sep. 14, 2012, now U.S. Pat. No. 8,565,339, issued on Oct. 22, 2013, which is a continuation of U.S. patent application Ser. No. 12/739,928, filed on Apr. 26, 2010, now U.S. Pat. No. 8,385,460, which is the national stage filing under 35 U.S.C. 371 of international application No. PCT/KR2008/007335, filed on Dec. 11, 2008, which claims the benefit of U.S. provisional application No. 61/012,889, filed on Dec. 11, 2007, and also claims the benefit of earlier filing date and right of priority to Korean patent application No. 10-2008-0124333, filed on Dec. 8, 2008, the contents of all of which are hereby incorporated by reference herein in their entireties.

TECHNICAL FIELD

The present invention relates to a method for transmitting and receiving a signal and an apparatus for transmitting and receiving a signal, and more particularly, to a method for transmitting and receiving a signal and an apparatus for transmitting and receiving a signal, which are capable of improving data transmission efficiency.

BACKGROUND ART

As a digital broadcasting technology has been developed, users have received a high definition (HD) moving image. With continuous development of a compression algorithm and high performance of hardware, a better environment will be provided to the users in the future. A digital television (DTV) system can receive a digital broadcasting signal and provide a variety of supplementary services to users as well as a video signal and an audio signal.

With the development of the digital broadcasting technology, a requirement for a service such as a video signal and an audio signal is increased and the size of data desired by a user or the number of broadcasting channels is gradually increased.

DISCLOSURE OF INVENTION Technical Problem

As a digital broadcasting technology has been developed, users have received a high definition (HD) moving image. With continuous development of a compression algorithm and high performance of hardware, a better environment will be provided to the users in the future. A digital television (DTV) system can receive a digital broadcasting signal and provide a variety of supplementary services to users as well as a video signal and an audio signal.

With the development of the digital broadcasting technology, a requirement for a service such as a video signal and an audio signal is increased and the size of data desired by a user or the number of broadcasting channels is gradually increased.

Technical Solution

An object of the present invention is to provide a method for transmitting and receiving a signal and an apparatus for transmitting and receiving a signal, which are capable of improving data transmission efficiency.

Another object of the present invention is to provide a method for transmitting and receiving a signal and an apparatus for transmitting and receiving a signal, which are capable of improving error correction capability of bits configuring a service.

Accordingly, the present invention is directed to a method for transmitting and receiving a signal and an apparatus for transmitting and receiving a signal that substantially obviate one or more problems due to limitations and disadvantages of the related art.

To achieve these objects and other advantages and in accordance with the purpose of the invention, as embodied and broadly described herein, a method for transmitting a signal is provided. The method includes converting (S110) a service stream to a physical layer pipe (PLP), allocating (S150) the PLP to a signal frame and arranging a preamble including a first pilot signal in a beginning part of the signal frame, converting (S160) the signal frame into a time domain according to an orthogonal frequency division multiplexing (OFDM) scheme, inserting (S170) a cyclic prefix obtained by frequency-shifting a first portion of an useful portion of the first pilot signal and a cyclic suffix obtained by frequency-shifting a second portion of said useful portion of the first pilot signal, into the first pilot signal and transmitting (S180) the signal frame including the first pilot signal over at least one radio frequency (RF) channel.

The first pilot signal may have a structure according to the following equation, B=one part (A)·e ^(j2πf) ^(SH) ^(t) C=another part (A)·e ^(j2πf) ^(SH) ^(t)

where, A denotes the valid portion of the first pilot signal, B denotes the cyclic prefix, C denotes the cyclic suffix, and fSH denotes a frequency shift unit.

The first portion may be a foremost portion of the useful portion of the first pilot signal and the second portion may be a backmost portion of the useful portion of the first pilot signal.

In another aspect of the present invention, a method for receiving a signal is provided. The method includes receiving (S210) the signal in a first frequency band, identifying (S220) a first pilot signal including a cyclic prefix obtained by frequency-shifting a first portion of an useful portion of the first pilot signal; and a cyclic suffix obtained by frequency-shifting a second portion of the useful portion of the first pilot signal from the received signal, demodulating (S220) a signal frame including a physical layer pipe (PLP) to which a service stream is converted, by an orthogonal frequency division multiplexing (OFDM) scheme, using information set in the first pilot signal, parsing (S230) the signal frame and obtaining the PLP, and obtaining (S240) the service stream from the PLP.

The demodulating of the signal frame includes estimating a timing offset and a fractional frequency offset of the received signal using the cyclic prefix and the cyclic suffix; and compensating for the estimated offsets.

In another aspect of the present invention, an apparatus for transmitting a signal is provided. The apparatus includes a coding and modulation unit (120) configured to error-correction-code a service stream and to interleave the error-correction-coded service stream, a frame builder (130) configured to map bits of the interleaved service stream to symbols of a physical layer pipe (PLP), allocate the PLP to a signal frame and arrange a preamble including a first pilot signal to a beginning part of the signal frame, a modulator (150 a) configured to convert the signal frame into a time domain according to an orthogonal frequency division multiplexing (OFDM) scheme and insert a cyclic prefix obtained by frequency-shifting a first portion of an useful portion of the first pilot signal and a cyclic suffix obtained by frequency-shifting a second portion of the useful portion of the first pilot signal, into the first pilot signal and a transmitter (160 a) configured to transmit the signal frame including the first pilot signal over at least one radio frequency (RF) channel.

In another aspect of the present invention, an apparatus for receiving a signal is provided. The apparatus includes first frequency band, a demodulator (220 a) configured to identify a first pilot signal including a cyclic prefix obtained by frequency-shifting a first portion of an useful portion of the first pilot signal and a cyclic suffix obtained by frequency-shifting a second portion of the useful portion of the first pilot signal from the received signal and demodulate a signal frame including a physical layer pipe (PLP), by an orthogonal frequency division multiplexing (OFDM) scheme, using information set in the first pilot signal, a frame parser (240) configured to parse the signal frame and obtaining the PLP and demap symbols of the PLP to bits of a service stream from the parsed signal frame and a decoding demodulator (250) configured to deinterleave the demapped bits of the service stream and decode the deinterleaved bits of the service stream using error-correction-decoding scheme.

In another aspect of the present invention, a method is provided. The method includes allocating (S150) a service stream to a signal frame and arranging a preamble including a pilot signal in a beginning part of the signal frame, modulating (S160) the signal frame, inserting (S170) a cyclic prefix obtained by modifying a first portion of an useful portion of the pilot signal and a cyclic suffix obtained by modifying a second portion of the useful portion of the pilot signal, into the pilot signal and transmitting (S180) the signal frame including the pilot signal.

In another aspect of the present invention, a method is provided. The method includes receiving (S210) the signal, identifying (S220) a signal frame from the received signal using a pilot signal including a cyclic prefix obtained by modifying a first portion of an useful portion of the pilot signal and a cyclic suffix obtained by modifying a second portion of the useful portion of first pilot signal and demodulating (S220) the signal frame, parsing (S230) the signal frame and obtaining (S240) a service stream from the parsed signal frame.

Advantageous Effects

According to the apparatus for transmitting and receiving the signal and the method for transmitting and receiving the signal of the invention, if the data symbol configuring the PLP and the symbols configuring the preamble are modulated in the same FFT mode, the probability that the data symbol is detected by the preamble is low and the probability that the preamble is erroneously detected is reduced. If continuous wave (CW) interference is included like the analog TV signal, the probability that the preamble is erroneously detected by a noise DC component generated at the time of correlation is reduced.

According to the apparatus for transmitting and receiving the signal and the method for transmitting and receiving the signal of the invention, if the size of the FFT applied to the data symbol configuring the PLP is larger than that of the FFT applied to the preamble, the preamble detecting performance may be improved even in a delay spread channel having a length equal to or greater than that of the useful symbol portion A of the preamble. Since both the cyclic prefix (B) and the cyclic suffix (C) are used in the preamble, the fractional carrier frequency offset can be estimated.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is a view showing a signal frame for transmitting a service;

FIG. 2 is a view showing the structure of a first pilot signal P1 of the signal frame;

FIG. 3 is a view showing a signaling window;

FIG. 4 is a schematic view showing an embodiment of an apparatus for transmitting a signal;

FIG. 5 is a view showing an example of an input processor 110;

FIG. 6 is a view showing an embodiment of a coding and modulation unit;

FIG. 7 is a view showing an embodiment of a frame builder;

FIG. 8 is a view showing a first example of a ratio of symbols when mappers 131 a and 131 b perform hybrid symbol mapping;

FIG. 9 is a view showing a second example of a ratio of symbols when the mappers 131 a and 131 b perform hybrid symbol mapping;

FIG. 10 is a view showing the number of symbols and bit number per cell word according to a symbol mapping scheme in an LDPC normal mode;

FIG. 11 is a view showing another example of the number of symbols according to a symbol mapping scheme in an LDPC normal mode;

FIG. 12 is a view showing another example of the number of symbols according to a symbol mapping scheme in an LDPC normal mode;

FIG. 13 is a view showing the number of symbols according to a symbol mapping scheme in an LDPC short mode;

FIG. 14 is a view showing an example of the number of symbols according to a symbol mapping scheme in an LDPC short mode;

FIG. 15 is a view showing another example of the number of symbols according to a symbol mapping scheme in an LDPC short mode;

FIG. 16 is a view showing an embodiment of each of the symbol mappers 131 a and 131 b shown in FIG. 7;

FIG. 17 is a view showing another embodiment of each of the symbol mappers 131 a and 131 b;

FIG. 18 is a view showing another embodiment of the symbol mapper;

FIG. 19 is a view showing another embodiment of each of the symbol mappers 131 a and 131 b;

FIG. 20 is a view showing the concept of interleaving of bits by bit interleavers 1312 a and 1312 b;

FIG. 21 is a view showing a first example of the number of rows and columns of memories of the bit interleavers 1312 a and 1312 b according to the types of symbol mappers 1315 a and 1315 b;

FIG. 22 is a view showing a second example of the number of rows and columns of the memories of the bit interleavers 1312 a and 1312 b according to the types of the symbol mappers 1315 a and 1315 b;

FIG. 23 is a diagram showing the concept of another embodiment of interleaving of a bit interleaver;

FIG. 24 is a view showing another embodiment of bit interleaving;

FIG. 25 is a view showing another embodiment of bit interleaving;

FIG. 26 is a view showing the concept of demultiplexing of input bits of demuxs 1313 a and 1313 b;

FIG. 27 is a view showing an embodiment of demultiplexing an input stream by the demux;

FIG. 28 is a view showing an example of a demultiplexing type according to a symbol mapping method;

FIG. 29 is a view showing an embodiment of demultiplexing an input bit stream according to a demultiplexing type;

FIG. 30 is a view showing a demultiplexing type which is determined according to a code rate of an error correction coding and a symbol mapping method;

FIG. 31 is a view showing an example of expressing the demultiplexing method by an equation;

FIG. 32 is a view showing an example of mapping a symbol by a symbol mapper;

FIG. 33 is a view showing an example of a multi-path signal coder;

FIG. 34 is a view showing an embodiment of a modulator;

FIG. 35 is a view showing an embodiment of an analog processor 160;

FIG. 36 is a view showing an embodiment of a signal receiving apparatus capable of receiving a signal frame;

FIG. 37 is a view showing an embodiment of a signal receiver;

FIG. 38 is a view showing an embodiment of a demodulator;

FIG. 39 is a view showing a multi-path signal decoder;

FIG. 40 is a view showing an embodiment of a frame parser;

FIG. 41 is a view showing an embodiment of each of symbol demappers 247 a and 247 p;

FIG. 42 is a view showing another embodiment of each of the symbol demappers 247 a and 247 p;

FIG. 43 is a view showing another embodiment of each of the symbol demappers 247 a and 247 p;

FIG. 44 is a view showing another embodiment of each of the symbol demappers 247 a and 247 p;

FIG. 45 is a view showing an embodiment of multiplexing a demultiplexed sub stream;

FIG. 46 is a view showing an example of a decoding and demodulation unit;

FIG. 47 is a view showing an embodiment of an output processor;

FIG. 48 is a view showing another embodiment of a signal transmitting apparatus for transmitting a signal frame;

FIG. 49 is a view showing another embodiment of a signal receiving apparatus for receiving a signal frame;

FIG. 50 is a view showing an embodiment of the structure of a first pilot signal;

FIG. 51 is a view showing an embodiment of detecting a preamble signal shown in FIG. 50 and estimating a timing offset and a frequency offset;

FIG. 52 is a view showing another embodiment of the structure of the first pilot signal;

FIG. 53 is a view showing an embodiment of detecting the first pilot signal shown in FIG. 52 and measuring a timing offset and a frequency offset;

FIG. 54 is a view showing an embodiment of detecting the first pilot signal and measuring a timing offset and a frequency offset using the detected result;

FIG. 55 is a view showing an embodiment of a method of transmitting a signal;

FIG. 56 is a view showing an embodiment of a method of receiving a signal; and

FIG. 57 is a flowchart illustrating an embodiment of identifying a first pilot signal and estimating an offset in a demodulating process.

BEST MODE FOR CARRYING OUT THE INVENTION

Reference will now be made in detail to the preferred embodiments of the present invention, examples of which are illustrated in the accompanying drawings. Wherever possible, the same reference numbers will be used throughout the drawings to refer to the same or like parts.

In the following description, the term “service” is indicative of either broadcast contents which can be transmitted/received by the signal transmission/reception apparatus, or content provision.

Prior to the description of an apparatus for transmitting and receiving a signal according to an embodiment of the present invention, a signal frame which is transmitted and received by the apparatus for transmitting and receiving the signal according to an embodiment of the present invention will be described.

FIG. 1 shows a signal frame for transmitting a service according to an embodiment of the present invention.

The signal frame shown in FIG. 1 shows an exemplary signal frame for transmitting a broadcast service including audio/video (A/V) streams. In this case, a single service is multiplexed in time- and frequency-channels, and the multiplexed service is transmitted. The above-mentioned signal transmission scheme is called a time-frequency slicing (TFS) scheme. Compared with the case in which a single service is transmitted to only one radio frequency (RF) band, the signal transmission apparatus according to an embodiment of the present invention transmits the signal service via at least one RF band (possibly several RF bands), such that it can acquire a statistical multiplexing gain capable of transmitting many more services. The signal transmission/reception apparatus transmits/receives a single service over several RF channels, such that it can acquire a frequency diversity gain.

First to third services (Services 1-3) are transmitted to four RF bands (RF1˜RF4). However, this number of RF bands and this number of services have been disclosed for only illustrative purposes, such that other numbers may also be used as necessary. Two reference signals (i.e., a first pilot signal (P1) and a second pilot signal (P2)) are located at the beginning part of the signal frame. For example, in the case of the RF1 band, the first pilot signal (P1) and the second pilot signal (P2) are located at the beginning part of the signal frame. The RF1 band includes three slots associated with the Service 1, two slots associated with the Service 2, and a single slot associated with the Service 3. Slots associated with other services may also be located in other slots (Slots 4˜17) located after the single slot associated with the Service 3.

The RF2 band includes a first pilot signal (P1), a second pilot signal (P2), and other slots 13˜17. In addition, the RF2 band includes three slots associated with the Service 1, two slots associated with the Service 2, and a single slot associated with the Service 3.

The Services 1˜3 are multiplexed, and are then transmitted to the RF3 and RF4 bands according to the time-frequency slicing (TFS) scheme. The modulation scheme for signal transmission may be based on an orthogonal frequency division multiplexing (OFDM) scheme.

In the signal frame, individual services are shifted to the RF bands (in the case that there is a plurality of the RF bands in the signal frame) and a time axis.

If signal frames equal to the above signal frame are successively arranged in time, a super-frame can be composed of several signal frames. A future extension frame may also be located among the several signal frames. If the future extension frame is located among the several signal frames, the super-frame may be terminated at the future extension frame.

FIG. 2 shows a first pilot signal (P1) contained in the signal frame of FIG. 1 according to an embodiment of the present invention.

The first pilot signal P1 and the second pilot signal P2 are located at the beginning part of the signal frame. The first pilot signal P1 is modulated by a 2K FFT mode, and may be transmitted simultaneously while including a ¼ guard interval. In FIG. 2, a band of 7.61 Mhz of the first pilot signal P1 includes a band of 6.82992 Mhz. The first pilot signal uses 256 carriers from among 1705 active carriers. A single active carrier is used for every 6 carriers on average. Data-carrier intervals may be irregularly arranged in the order of 3, 6, and 9. In FIG. 2, a solid line indicates the location of a used carrier, a thin dotted line indicates the location of an unused carrier, and a chain line indicates a center location of the unused carrier. In the first pilot signal, the used carrier can be symbol-mapped by a binary phase shift keying (BPSK), and a pseudo-random bit sequence (PRBS) can be modulated. The size of a FFT used for the second pilot signal can be indicated by several PRBSs.

The signal reception apparatus detects a structure of a pilot signal, and recognizes a time-frequency slicing (TFS) using the detected structure. The signal reception apparatus acquires the FFT size of the second pilot signal, compensates for a coarse frequency offset of a reception signal, and acquires time synchronization.

In the first pilot signal, a signal transmission type and a transmission parameter may be set.

The second pilot signal P2 may be transmitted with a FFT size and a guard interval equal to those of the data symbol. In the second pilot signal, a single carrier is used as a pilot carrier at intervals of three carriers. The signal reception apparatus compensates for a fine frequency synchronization offset using the second pilot signal, and performs fine time synchronization. The second pilot signal transmits information of a first layer (L1) from among Open Systems Interconnection (OSI) layers. For example, the second pilot signal may include a physical parameter and frame construction information. The second pilot signal transmits a parameter value by which a receiver can access a Physical Layer Pipe (PLP) service stream.

L1 (Layer 1) information contained in the second pilot signal P2 is as follows.

The Layer-1 (L1) information includes a length indicator indicating the length of data including the L1 information, such that it can easily use the signaling channels of Layers 1 and 2 (L1 and L2). The Layer-1 (L1) information includes a frequency indicator, a guard-interval length, a maximum number of FEC (Forward Error Correction) blocks for each frame in association with individual physical channels, and the number of actual FEC blocks to be contained in the FEC block buffer associated with a current/previous frame in each physical channel. In this case, the frequency indicator indicates frequency information corresponding to the RF channel.

The Layer-1 (L1) information may include a variety of information in association with individual slots. For example, the Layer-1 (L1) information includes the number of frames associated with a service, a start address of a slot having the accuracy of an OFDM carrier contained in an OFDM symbol, a length of the slot, slots corresponding to the OFDM carrier, the number of bits padded in the last OFDM carrier, service modulation information, service mode rate information, and Multi-Input-Multi-Output (MIMO) scheme information.

The Layer-1 (L1) information may include a cell ID, a flag for service like notification message service (e.g., an emergency message), the number of current frames, and the number of additional bits for future use. In this case, the cell ID indicates a broadcast area transmitted by a broadcast transmitter

The second pilot signal P2 is adapted to perform channel estimation for decoding a symbol contained in the P2 signal. The second pilot signal P2 can be used as an initial value for channel estimation for the next data symbol. The second pilot signal P2 may also transmit Layer-2 (L2) information. For example, the second pilot signal is able to describe information associated with the transmission service in Layer-2 (L2) information. The signal transmission apparatus decodes the second pilot signal, such that it can acquire service information contained in the time-frequency slicing (TFS) frame and can effectively perform the channel scanning. Meanwhile, this Layer-2 (L2) information may be included in a specific PLP of the TFS frame. According to another instance, L2 information can be included in a specific PLP, and the service description information also can be transmitted in the specific PLP.

For example, the second pilot signal may include two OFDM symbols of the 8k FFT mode. Generally, the second pilot signal may be any one of a single OFDM symbol of the 32K FFT mode, a single OFDM symbol of the 16k FFT mode, two OFDM symbols of the 8k FFT mode, four OFDM symbols of the 4k FFT mode, and eight OFDM symbols of the 2k FFT mode.

In other words, a single OFDM symbol having the size of a large FFT or several OFDM symbols, each of which has the size of a small FFT, may be contained in the second pilot signal P2 such that capacity capable of being transmitted to the pilot can be maintained.

If information to be transmitted to the second pilot signal exceeds capacity of the OFDM symbol of the second pilot signal, OFDM symbols after the second pilot signal can be further used. L1 (Layer1) and L2 (Layer2) information contained in the second pilot signal is error-correction-coded and is then interleaved, such that data recovery is carried out although an impulse noise occurs.

As described the above, L2 information can also be included in a specific PLP conveying the service description information.

FIG. 3 shows a signaling window according to an embodiment of the present invention. The time-frequency slicing (TFS) frame shows an offset concept of the signaling information. Layer-1 (L1) information contained in the second pilot signal includes frame construction information and physical layer information required by the signal reception apparatus decoding the data symbol. Therefore, if information of the following data symbols located after the second pilot signal, is contained in the second pilot signal, and the resultant second pilot signal is transmitted, the signal reception apparatus may be unable to immediately decode the above following data symbols due to a decoding time of the second pilot signal.

Therefore, as shown in FIG. 3, the L1 information contained in the second pilot signal (P2) includes information of a single time-frequency slicing (TFS) frame size, and includes information contained in the signaling window at a location spaced apart from the second pilot signal by the signaling window offset.

In the meantime, in order to perform channel estimation of a data symbol constructing the service, the data symbol may include a scatter pilot and a continual pilot.

The signal transmission/reception system capable of transmitting/receiving signal frames shown in FIGS. 1˜3 will hereinafter be described. Individual services can be transmitted and received over several RF channels. A path for transmitting each of the services or a stream transmitted via this path is called a PLP. The PLP may be distributed among the timely-divided slots in several RF channels or a single RF band. This signal frame can convey the timely-divided PLP in at least one RF channel. In other word, a single PLP can be transferred through at least one RF channel with timely-divided regions. Hereinafter the signal transmission/reception systems transmitting/receiving a signal frame via at least one RF band will be disclosed.

FIG. 4 is a block diagram illustrating an apparatus for transmitting a signal according to one embodiment of the present invention. Referring to FIG. 4, the signal transmission apparatus includes an input processor 110, a coding and modulation unit 120, a frame builder 130, a MIMO/MISO encoder 140, a plurality of modulators (150 a, . . . , 150 r) of the MIMO/MISO encoder 140, and a plurality of analog processors (160 a, . . . , 160 r).

The input processor 110 receives streams equipped with several services, generates P number of baseband frames (P is a natural number) which includes modulation- and coding-information corresponding to transmission paths of the individual services, and outputs the P number of baseband frames.

The coding and modulation unit 120 receives baseband frames from the input processor 110, performs the channel coding and interleaving on each of the baseband frames, and outputs the channel coding and interleaving result.

The frame builder 130 forms frames which transmit baseband frames contained in P number of PLPs to R number of RF channels (where R is a natural number), splits the formed frames, and outputs the split frames to paths corresponding to the R number of RF channels. Several services may be multiplexed in a single RF channel in time. The signal frames generated from the frame builder 140 may include a time-frequency slicing (TFS) structure in which the service is multiplexed in time- and frequency-domains.

The MIMO/MISO encoder 140 encodes signals to be transmitted to the R number of RF channels, and outputs the coded signals to paths corresponding to A number of antennas (where A is a natural number). The MIMO/MISO encoder 140 outputs the coded signal in which a single to be transmitted to a single RF channel is encoded to the A number of antennas, such that a signal is transmitted/received to/from a MIMO (Multi-Input-Multi-Output) or MISO (Multi-Input-Single-Output) structure.

The modulators (150 a, . . . , 150 r) modulate frequency-domain signals entered via the path corresponding to each RF channel into time-domain signals. The modulators (150 a, . . . , 150 r) modulate the input signals according to an orthogonal frequency division multiplexing (OFDM) scheme, and outputs the modulated signals.

The analog processors (160 a, . . . , 160 r) convert the input signals into RF signals, such that the RF signals can be outputted to the RF channels.

The signal transmission apparatus according to this embodiment may include a predetermined number of modulators (150 a, . . . 150 r) corresponding to the number of RF channels and a predetermined number of analog processors (160 a, . . . , 160 r) corresponding to the number of RF channels. However, in the case of using the MIMO scheme, the number of analog processors must be equal to the product of R (i.e., the number of RF channels) and A (i.e., the number of antennas).

FIG. 5 is a block diagram illustrating an input processor 110 according to an embodiment of the present invention. Referring to FIG. 5, the input processor 110 includes the first stream multiplexer 111 a, the first service splitter 113 a, and a plurality of first baseband (BB) frame builders (115 a, . . . , 115 m). The input processor 110 includes a second stream multiplexer 111 b, a second service splitter 113 b, and a plurality of second baseband (BB) frame builders (115 n, . . . , 115 p).

For example, the first stream multiplexer 111 a receives several MPEG-2 transport streams (TSs), multiplexes the received MPEG-2 TS streams, and outputs the multiplexed MPEG-2 TS streams. The first service splitter 113 a receives the multiplexed streams, splits the input streams of individual services, and outputs the split streams. As described above, provided that the service transmitted via a physical-channel path is called a PLP, the first service splitter 113 a splits the service to be transmitted to each PLP, and outputs the split service.

The first BB frame builders (115 a, . . . , 115 m) build data contained in a service to be transmitted to each PLP in the form of a specific frame, and output the specific-frame-formatted data. The first BB frame builders (115 a, . . . , 115 m) build a frame including a header and payload equipped with service data. The header of each frame may include mode information based on the modulation and encoding of the service data, and a counter value based on a clock rate of the modulator to synchronize input streams.

The second stream multiplexer 111 b receives several streams, multiplexes input streams, and outputs the multiplexed streams. For example, the second stream multiplexer 111 b may multiplex Internet Protocol (IP) streams instead of the MPEG-2 TS streams. These streams may be encapsulated by a generic stream encapsulation (GSE) scheme. The streams multiplexed by the second stream multiplexer 111 b may be any one of streams. Therefore, the above-mentioned streams different from the MPEG-2 TS streams are called generic streams (GS streams).

The second service splitter 113 b receives the multiplexed generic streams, splits the received generic streams according to individual services (i.e., PLP types), and outputs the split GS streams.

The second BB frame builders (115 n, . . . , 115 p) build service data to be transmitted to individual PLPs in the form of a specific frame used as a signal processing unit, and output the resultant service data. The frame format built by the second BB frame builders (115 n, . . . , 115 p) may be equal to that of the first BB frame builders (115 a, . . . , 115 m) as necessary. If required, another embodiment may also be proposed. In another embodiment, the frame format built by the second BB frame builders (115 n, . . . , 115 p) may be different from that of the first BB frame builders (115 a, . . . , 115 m). The MPEG-2 TS header further includes a Packet Syncword which is not contained in the GS stream, resulting in the occurrence of different headers.

FIG. 6 is a block diagram illustrating a coding and modulation unit according to an embodiment of the present invention. The coding and modulation unit includes a first interleaver 123, a second encoder 125, and a second interleaver 127.

The first encoder 121 acts as an outer coder of the input baseband frame, and is able to perform the error correction encoding. The first encoder 121 performs the error correction encoding of the input baseband frame using a Bose-Chaudhuri-Hocquenghem (BCH) scheme. The first interleaver 123 performs interleaving of the encoded data, such that it prevents a burst error from being generated in a transmission signal. The first interleaver 123 may not be contained in the above-mentioned embodiment.

The second encoder 125 acts as an inner coder of either the output data of the first encoder 121 or the output data of the first interleaver 123, and is able to perform the error correction encoding. A low density parity bit (LDPC) scheme may be used as an error correction encoding scheme. The second interleaver 127 mixes the error-correction-encoded data generated from the second encoder 125, and outputs the mixed data. The first interleaver 123 and the second interleaver 127 are able to perform interleaving of data in units of a bit.

The coding and modulation unit 120 relates to a single PLP stream. The PLP stream is error-correction-encoded and modulated by the coding and modulation unit 120, and is then transmitted to the frame builder 130.

FIG. 7 is a block diagram illustrating a frame builder according to an embodiment of the present invention. Referring to FIG. 7, the frame builder 130 receives streams of several paths from the coding and modulation unit 120, and arranges the received streams in a single signal frame. For example, the frame builder may include a first mapper 131 a and a first time interleaver 132 a in a first path, and may include a second mapper 131 b and a second time interleaver 132 b in a second path. The number of input paths is equal to the number of PLPs for service transmission or the number of streams transmitted via each PLP.

The first mapper 131 a performs mapping of data contained in the input stream according to the first symbol mapping scheme. For example, the first mapper 131 a may perform mapping of the input data using a QAM scheme (e.g., 16 QAM, 64 QAM, and 256 QAM).

If the first mapper 131 a performs mapping of the symbol, the input data may be mapped to several kinds of symbols according to several symbol mapping schemes. For example, the first mapper 131 a classifies the input data into a baseband-frame unit and a baseband-frame sub-unit. Individual classified data may be hybrid-symbol-mapped by at least two QAM schemes (e.g., 16 QAM and 64 QAM). Therefore, data contained in a single service may be mapped to symbols based on different symbol mapping schemes in individual intervals.

The first time interleaver 132 a receives a symbol sequence mapped by the first mapper 131 a, and is able to perform the interleaving in a time domain. The first mapper 131 a maps data, which is contained in the error-corrected frame unit received from the coding and modulation unit 120, into symbols. The first time interleaver 132 a receives the symbol sequence mapped by the first mapper 131 a, and interleaves the received symbol sequence in units of the error-corrected frame.

In this way, the p-th mapper 131 p or the p-th time interleaver 132 p receives service data to be transmitted to the p-th PLP, maps the service data into symbols according to the p-th symbol mapping scheme. The mapped symbols can be interleaved in a time domain. It should be noted that this symbol mapping scheme and this interleaving scheme are equal to those of the first time interleaver 132 a and the first mapper 131 a.

The symbol mapping scheme of the first mapper 131 a may be equal to or different from that of the p-th mapper 131 p. The first mapper 131 a and the p-th mapper 131 p are able to map input data to individual symbols using the same or different hybrid symbol mapping schemes.

Data of the time interleavers located at individual paths (i.e., service data interleaved by the first time interleaver 132 a and service data to be transmitted to R number of RF channels by the p-th time interleaver 132 p) is interleaved, such that the physical channel allows the above data to be interleaved over several RF channels.

In association with streams received in as many paths as the number of PLPs, the TFS frame builder 133 builds the TFS signal frame such as the above-mentioned signal frame, such that the service is time-shifted according to RF channels. The TFS frame builder 133 splits service data received in any one of paths, and outputs the service data split into data of the R number of RF bands according to a signal scheduling scheme.

The TFS frame builder 133 receives the first pilot signal and the second pilot signal from the signaling information unit (denoted by Ref/PL signal) 135, arranges the first and second pilot signals in the signal frame, and inserts the signaling signal (L1 and L2) of the above-mentioned physical layer in the second pilot signal. In this case, the first and second pilot signals are used as the beginning signals of the signal frame contained in each RF channel from among the TFS signal frame received from the signaling information unit (Ref/PL signal) 135. As shown in FIG. 2, the first pilot signal may include a transmission type and basic transmission parameters, and the second pilot signal may include a physical parameter and frame construction information. Also, the second pilot signal includes a L1 (Layer 1) signaling signal and a L2 (Layer 2) signaling signal.

The R number of frequency interleavers (137 a, . . . , 137 r) interleave service data, to be transmitted to corresponding RF channels of the TFS signal frame, in a frequency domain. The frequency interleavers (137 a, . . . , 137 r) can interleave the service data at a level of data cells contained in an OFDM symbol.

Therefore, service data to be transmitted to each RF channel in the TFS signal frame is frequency-selective-fading-processed, such that it may not be lost in a specific frequency domain.

FIG. 8 is a view showing a first example of a ratio of symbols when the mappers 131 a and 131 b perform hybrid symbol mapping. This Figure shows the number of bits transmitted by one sub carrier (cell) if error correction coding is performed by the coding and modulation unit in a normal mode (the length of the error-correction-coded code is 64800 bits) of LDPC error correction coding mode.

For example, if the mappers 131 a and 131 b perform symbol mapping using 256 QAM, 64800 bits are mapped to 8100 symbols. If the mappers 131 a and 131 b perform hybrid symbol mapping (Hyb 128-QAM) using 256 QAM and 64 QAM with a ratio of 3:2, the number of symbols mapped by 256 QAM is 4860 and the number of symbols mapped by 64 QAM is 4320. The number of transmitted bits per sub carrier (cell) is 7.0588.

If a symbol mapping method of 64 QAM is used, input data may be mapped to 10800 symbols and six bits per cell may be transmitted. If data is mapped to the symbols by a hybrid symbol mapping method of 64 QAM and 16 QAM (64 QAM:16 QAM=3:2, Hyb32-QAM), five bits may be transmitted by one sub carrier (cell).

If data is mapped to symbols by the 16 QAM method, the data is mapped to 16200 symbols, each of which is used to transmit four bits.

Similarly, if data is mapped to symbols by a hybrid symbol mapping method of 16 QAM and QPSK (16 QAM:QPSK=2:3, Hyb8-QAM), three bits may be transmitted by one sub carrier (cell).

If data is mapped to symbols by a QPSK method, the data may be mapped to 32400 symbols, each of which is used to transmit two bits.

FIG. 9 shows symbol mapping methods of error-corrected data by LDPC error correction coding method of a short mode (the length of the error-correction-coded code is 16200 bits), which are equal to the symbol mapping methods of FIG. 8, and the numbers of bits per sub carrier according to the symbol mapping methods.

The numbers of bits transmitted by the sub carrier is equal to those of the normal mode (64800 bits) according to the symbol mapping methods such as 256 QAM, Hyb 128-QAM, 64-QAM, Hyb 32 -QAM, 16 QAM, Hyb8-QAM and QPSK, but the total numbers of symbols transmitted are different from those of the normal mode. For example, 16200 bits are transmitted by 2025 symbols in 256 QAM, 16200 bits are transmitted by 1215 symbols according to 256 QAM and 1080 symbols according to 64 QAM (total 2295 symbols) in Hyb 128-QAM.

Accordingly, a data transmission rate per sub carrier (cell) for each PLP may be adjusted according to a hybrid symbol mapping method or a single symbol mapping method.

FIG. 10 is a view showing the number of symbols and bit number per cell word according to a symbol mapping method in an LDPC normal mode. If a TFS signal frame includes at least one RF channel, symbols configuring a specific PLP can be uniformly allocated to RF channels. The locations of the PLP symbols allocated to the RF channels can be more efficiently addressed. Accordingly, when the signal receiving apparatus selects the RF channels, the bits used for addressing the specific PLP can be reduced.

In this drawing, a symbol mapping method represented by 256-QAM indicates a method of mapping bits configuring a single error-correction-coded block to symbols with a ratio of 256 QAM:64 QAM=8:1. According to this symbol mapping method, the number of the bits in a single error-correction-coded block by the 256-QAM method is 57600, the number of the bits in a single error-correction-coded block by the 256-QAM method is 1200, the number of total symbols in the block is 8400, and the bit number per cell word is 7.714285714.

A symbol mapping method represented by Hyb 128-QAM indicates a method of mapping bits configuring a single error-correction-coded block to symbols with a ratio of 256 QAM:64 QAM=8:7. According to the Hyb 128-QAM symbol mapping method, the number of total symbols in a single error-correction-encoding block is 9600, and the bit number per cell word is 6.75.

According to a symbol mapping method represented by 64 QAM, the number of total symbols in a single error-correction-encoding block is 10800 and the bit number per cell word is 6.

A symbol mapping method represented by Hyb 32-QAM indicates a method of mapping bits configuring a single error-correction-coded block to symbols with a ratio of 64 QAM:32 QAM=5:4. According to the Hyb 32-QAM symbol mapping method, the number of total symbols in the error-correction-coded block is 13200, and the bit number per cell word is 4.9090909.

A symbol mapping method represented by 16 QAM indicates a method of mapping bits configuring a single error-correction-coded block to symbols with a ratio of 16 QAM:QPSK=1:8. According to the 16 QAM symbol mapping method, the number of total symbols in one error-correction-coded block is 15600, and the bit number per cell word is 4.153846154.

A symbol mapping method represented by Hyb 8-QAM indicates a method of mapping bits configuring a single error-correction-coded block to symbols with a ratio of 16 QAM:QPSK=2:1. According to the Hyb 8-QAM symbol mapping method, the number of total symbols in one error-correction-coded block is 21600, and the bit number per cell word is 3.

According to a symbol mapping method represented by QPSK, the number of total symbols in one error-correction-coded block is 32400 and the bit number per cell word is 2.

When the symbols configuring the PLP are allocated to the RF channels, the diversity gain of the frequency domain can be maximized when the numbers of the symbols allocated to the respective RF channels are equal. If a maximum of six RF channels is considered, the lowest common multiple of 1 to 6 is 60 and the greatest common divisor of the numbers of symbols mapped to one error correction coded block is 1200. Accordingly, if the integral multiple of 1200/60=20 symbols is allocated to each of the RF channels, the symbols can be uniformly allocated to all the RF channels. At this time, if 20 symbols are considered as one group and the group is addressed, the addressing overhead of log2(20)≈4.32 bits can be reduced compared with the case the symbols are addressed one by one.

FIG. 11 is a view showing another example of the number of symbols according to a symbol mapping method in an LDPC normal mode. In the example of this drawing, a 256-QAM method using 256 QAM and 64 QAM symbols (256 QAM:64 QAM=4:1), a Hyb 128-QAM method using 256 QAM and 64 QAM symbol (256 QAM:64 QAM=8:7), a 64 QAM method, a Hyb 32-QAM method using 64 QAM and 8 QAM symbols (64 QAM:8 QAM=3:2), a 16 QAM method using 16 QAM and QPSK symbols (16 QAM:QPSK=1:14), a Hyb 8-QAM method using 16 QAM:QPSK=2:1 and a QPSK method were used as the symbol mapping method. The greatest common divisor (GCD) of the numbers of total symbols of an error correction coded block (normal mode) according to the symbol mapping methods is 720. Accordingly, if the integral multiple of 12(=720/60) symbols is allocated to each of the RF channels, the symbols can be uniformly allocated to all the RF channels. At this time, if 12 symbols are considered as one group and the group is addressed, the addressing overhead of log2(12)≈3.58 bits can be reduced compared with the case the symbols are addressed one by one. The signal receiving apparatus can collect the allocated PLP symbols by the addressing scheme and obtain a PLP service stream.

FIG. 12 is a view showing another example of the number of symbols according to a symbol mapping method in an LDPC normal mode. In the example of this drawing, a 256-QAM scheme, a Hyb 128-QAM scheme, a 64 QAM scheme, a Hyb 32-QAM scheme, a 16 QAM scheme, a Hyb 8-QAM scheme and a QPSK scheme were used as the symbol mapping method. The 256 QAM symbol mapping method uses 256 QAM and 64 QAM symbols (256 QAM:64 QAM=44:1) and the Hyb 128-QAM symbol mapping method uses 256 QAM and 64 QAM symbols (256 QAM:64 QAM=28:17). The Hyb 32-QAM method uses 64 QAM and 8 QAM symbols (64 QAM:8 QAM=3:2), the 16 QAM symbol mapping method uses 16 QAM and QPSK symbols (16 QAM:QPSK=1:14), and the Hyb 8-QAM symbol mapping method uses 16 QAM and QPSK symbols (16 QAM:QPSK=2:1). The GCD of the numbers of total symbols of an error correction coded block (normal mode) according to the symbol mapping methods is 240. Accordingly, if the integral multiple of 240/60=4 symbols is allocated to each of the RF channels, the symbols can be uniformly allocated to all the RF channels. At this time, if four symbols are considered as one group and the group is addressed, the addressing overhead of log2(4)≈2 bits can be reduced compared with the case where the symbols are addressed one by one. Accordingly, even when the number of RF channels is any one of 1 to 6 in the signal frame, the PLP symbols can be uniformly allocated to the RF channels.

FIG. 13 is a view showing the number of symbols according to a symbol mapping method in an LDPC short mode. As described above, if symbol mapping is performed according to this example, the PLP symbols can be uniformly allocated to the RF channels and the overhead of the PLP symbol addressing can be reduced. The symbol mapping methods shown in this drawing are equal to those shown in FIG. 10. However, since the bit number of the LDPC short mode is different from that of the normal mode, the GCD of the numbers of total symbols of an error correction coded block (short mode) according to the symbol mapping methods is 300, unlike to FIG. 10. Accordingly, if the integral multiple of 300/60=5 symbols is allocated to each of the RF channels, the symbols can be uniformly allocated to all the RF channels. At this time, if five symbols are considered as one group and the group is addressed, the addressing overhead of log2(5) bits can be reduced compared with the case where the symbols are addressed one by one. Accordingly, in this embodiment, the addressing bits are saved by log2(5) bits when the divided PLP symbols are addressed.

FIG. 14 is a view showing an example of the number of symbols according to a symbol mapping method in an LDPC short mode. The symbol mapping methods of this drawing are equal to those shown in FIG. 11. In this example, the GCD of the numbers of total symbols of an error correction coded block (short mode) according to the symbol mapping methods is 180, which may be used for PLP symbol allocation of one RF channel and the addressing of the allocated symbols. In this embodiment, the addressing bits are saved by log2(3) bits.

FIG. 15 is a view showing another example of the number of symbols according to a symbol mapping method in an LDPC short mode. The symbol mapping methods of this drawing are equal to those shown in FIG. 12. In this example, the GCD of the numbers of total symbols of an error correction coded block (short mode) according to the symbol mapping methods is 60. In this embodiment, the addressing bits are saved by log2(1) bits (that is, the addressing bit is not saved).

FIG. 16 is a view showing an example of each of the symbol mappers 131 a and 131 b shown in FIG. 7. Each of the symbol mappers 131 a and 131 b includes a first order mapper 1315 a, a second order mapper 131 b, a symbol merger 1317 and an error correction block merger 1318.

The bit stream parser 1311 receives the PLP service stream from the coding and modulation unit and splits the received service stream.

The first order symbol mapper 1315 a maps the bits of the service stream split by a higher order symbol mapping method to symbols. The second order symbol mapper 1315 b maps the bits of the service stream split by a lower order symbol mapping method to symbols. For example, in the above example, the first order symbol mapper 1315 a may map the bit stream to symbols according to 256 QAM and the second order symbol mapper 1315 b may map the bit stream to symbols according to 64 QAM.

The symbol merger 1317 merges the symbols output from the symbol mappers 1315 a and 1315 b to one symbol stream and outputs the symbol stream. The symbol merger 1317 may output the symbol stream included in one PLP.

The error correction block merger 1318 may output one symbol stream merged by the symbol merger 1317 in the error-correction-coded code block unit. The error correction block merger 1318 may output a symbol block such that the error-correction-coded code blocks are uniformly allocated to at least one RF band of the TFS signal frame. The error correction block merger 1318 may output the symbol block such that the length of the symbol block of the error-correction-coded block of a normal mode is equal to that of the symbol block of the error-correction-coded block of a short mode. For example, four symbol blocks of the error-correction-coded block of the short mode may be merged to one symbol block.

The error correction block merger 1318 may split the symbol stream according to a common multiple of the number of RF bands such that signal frame builder uniformly arranges the symbols to the RF bands. If the maximum number of RF bands in the signal frame is 6, the error correction block merger 1318 outputs the symbol block such that the total number of symbols can be divided by 60 which is a common multiple of 1, 2, 3, 4, 5 and 6.

The symbols included in the output symbol block may be arranged to be uniformly allocated to the six RF bands. Accordingly, although an error correction mode according to a code rate and a symbol mapping method are combined, the symbols configuring the PLP are uniformly allocated to the RF bands.

FIG. 17 is a view showing another embodiment of each of the symbol mappers 131 a and 131 b. The embodiment of this drawing is similar to the embodiment of FIG. 16 except that a first order power calibration unit 1316 a and a second order power calibration unit 1316 b are further included.

The first order power calibration unit 1316 a calibrates the power of the symbols mapped by the first order symbol mapper 1315 a according to the size of the constellation and outputs the calibrated symbols. The second order power calibration unit 1316 b calibrates the power of the symbols mapped by the second order symbol mapper 1315 b according to the size of the constellation and outputs the calibrated symbols. Accordingly, although the symbol mapping method is changed in one PLP or is changed among a plurality of PLPs, if the power of the symbol by the symbol mapping method is adjusted according to the size of the constellation, signal reception performance of a receiver can be improved.

The symbol merger 1317 merges the symbols calibrated by the power calibration units 1316 a and 1316 b and outputs one symbol stream.

FIG. 18 is a view showing another embodiment of the symbol mapper. In the embodiment of this Figure, the symbol mapper includes the second encoder 125 and the second interleaver 127 included in the coding and modulation unit. That is, if this embodiment is used, the coding and modulation unit may include only the first encoder 121, the first interleaver 123 and the second encoder 125.

The embodiment of the symbol mapper includes a bit stream parser 1311, a first order bit interleaver 1312 a, a second order bit interleaver 1312 b, a first order demux 1313 a, a second order demux 1313 b, a first order symbol mapper 1315 a, a second order symbol mapper 1315 b and a symbol merger 1317.

When the second encoder 125 performs LDPC error correction coding, the length of the error-correction-coded block (e.g., the length of 64800 bits and the length of 16200 bits) may vary according to an LDPC mode. If the bits included in the error-correction-coded block are mapped to the symbols, the error correction capabilities of the bits included in a cell word configuring the symbol may vary according to the locations of the bits. For example, the cell word which is the symbol may be determined according to the code rate of the error correction coding and the symbol mapping method (whether the symbol mapping method is the higher order symbol mapping method or the lower order symbol mapping method). If the error-correction-code is the LDPC, the error correction capabilities of the bits vary according to the locations of the bits in the error-correction-coded block. For example, the reliabilities of the bits coded according to the characteristics of the H-matrix used in the irregular LDPC error correction coding method may vary according to the locations of the bits. Accordingly, the order of the bits configuring the cell word mapped to the symbol is changed such that the error correction capabilities of the bits which are weak against the error correction in the error-correction-coded block are adjusted and the robustness against the error in the bit level can be adjusted.

First, the second encoder 125, for example, performs the error correction coding with respect to the stream included in one PLP by the LDPC error correction coding method.

The bit stream parser 1311 receives the service stream according to the PLP and splits the received service stream.

The first order bit interleaver 1312 a interleaves the bits included in a first bit stream of the split service streams. Similarly, the second order bit interleaver 1312 b interleaves the bits included in a second bit stream of the split service streams.

The first order bit interleaver 1312 a and the second order bit interleaver 1312 b may correspond to the second interleaver 127 used as an inner interleaver. The interleaving method of the first order bit interleaver 1312 a and the second order bit interleaver 1312 b will be described later.

The first order demux 1313 a and the second order demux 1313 b demultiplex the bits of the bit streams interleaved by the first order bit interleaver 1312 a and the second order bit interleaver 1312 b. The demuxs 1313 a and 1313 b divide the input bit stream into sub bit streams which will be mapped to a real axis and an imaginary axis of a constellation and output the sub bit streams. The symbol mappers 1315 a and 1315 b map the sub bit streams demultiplexed by the demuxs 1313 a and 1313 b to the corresponding symbols.

The bit interleavers 1312 a and 1312 b and the demuxs 1313 a and 1313 b may combine the characteristics of the LDPC codeword and the characteristics of the constellation reliability of the symbol mapping according to the constellation. The detailed embodiment of the first order demuxs 1313 a and 1313 b will be described later.

The first order symbol mapper 1315 a performs first order symbol mapping, for example, higher order symbol mapping, and the second order symbol mapper 1315 b performs second order symbol mapping, for example, lower order symbol mapping. The first order symbol mapper 1315 a maps the sub bit streams output from the first order demux 1313 to the symbols and the second order symbol mapper 1315 b maps the sub bit streams output from the second order demux 1313 b to the symbols.

The symbol merger 1317 merges the symbols mapped by the first order symbol mapper 1315 a and the second order symbol mapper 1315 b to one symbol stream and outputs the symbol stream.

As described above, in the LDPC, the error correction capabilities of the bits may be changed according to the locations of the bits in the error-correction-coded block. Accordingly, if the bit interleaver and the demux are controlled according to the characteristics of the LDPC encoder 125 so as to change the order of the bits configuring the cell word, the error correction capability in the bit level can be maximized.

FIG. 19 is a view showing another embodiment of each of the symbol mappers 131 a and 131 b. The embodiment of this drawing is similar to the embodiment of FIG. 18 except that a first order power calibration unit 1316 a and a second order power calibration unit 1316 b are further included.

The first order power calibration unit 1316 a calibrates the power of the symbols mapped by the first order symbol mapper 1315 a according to the size of the constellation and outputs the calibrated symbols. The second order power calibration unit 1316 b calibrates the power of the symbols mapped by the second order symbol mapper 1315 b according to the size of the constellation and outputs the calibrated symbols. Accordingly, although the symbol mapping scheme is changed in one PLP or is changed among a plurality of PLPs, if the power of the symbol is adjusted according to the size of the constellation, signal reception performance can be improved.

The symbol merger 1317 merges the symbols calibrated by the power calibration units 1316 a and 1316 b and outputs one symbol stream.

FIG. 20 is a view showing the concept of interleaving of bits by the bit interleavers 1312 a and 1312 b of FIGS. 18 and 19.

For example, input bits are stored in and read from a matrix-formed memory having a predetermined number of rows and columns. When the input bits are stored, first, the bits are stored in a first column in row direction, and, if the first column is filled up, the bits are stored in another column in row direction. When the stored bits are read, the bits are read in column direction and, if all the bits stored in a first row are read, the bits in another row are read in column direction. In other word, when the bits are stored, the bits are stored row-wise such that the columns are filled up serially. And when the stored bits are read, the stored bits are read column-wise from the first row to last row serially. In this Figure, MSB means a most significant bit and LSB means a least significant bit.

In order to map the LDPC-error-correction-coded bits to the symbols in the same length of error correction block unit at various code rates, the bit interleavers 1312 a and 1312 b may change the number of rows and columns of the memory according to the types of the symbol mappers 1315 a and 1315 b.

FIG. 21 is a view showing an example of the number of rows and columns of memories of the bit interleavers 1312 a and 1312 b according to the types of symbol mappers 1315 a and 1315 b, if the LDPC mode is the normal mode.

For example, if the symbol mapper 1315 a maps the bits to 256 QAM symbols, the first order interleaver 1312 a interleaves the bits by a memory having 8100 rows and 8 columns. If the symbols are mapped by 64 QAM, the first order interleaver 1312 a interleaves the bits by a memory having 10800 rows and 6 columns. If the symbols are mapped by 16 QAM, the first order interleaver 1312 a interleaves the bits by a memory having 16200 rows and 4 columns.

For example, if the symbol mappers 1315 a and 1315 b map the bits to Hyb128-QAM symbols, the first order interleaver 1312 a interleaves the bits using a memory having 4860 rows and 8 columns, and the second order interleaver 1312 b interleaves the bits using a memory having 4320 rows and 6 columns.

Similarly, if the symbol mappers 1315 a and 1315 b map the symbols by Hyb32-QAM, the first order interleaver 1312 a interleaves the bits using a memory having 6480 rows and 6 columns, and the second order interleaver 1312 b interleaves the bits using a memory having 6480 rows and 4 columns.

FIG. 22 is a view showing an example of the number of rows and columns of the memories of the bit interleavers 1312 a and 1312 b according to the types of the symbol mappers 1315 a and 1315 b, if the LDPC mode is the short mode.

For example, if the symbol mapper 1315 a maps the bits to 256 QAM symbols, the first order interleaver 1312 a interleaves the bits by a memory having 2025 rows and 8 columns. If the symbol mappers 1315 a and 1315 b map the symbols by Hyb128-QAM, the first order interleaver 1312 a interleaves the bits using a memory having 1215 rows and 8 columns, and the second order interleaver 1312 b interleaves the bits using a memory having 1080 rows and 6 columns.

If the bit interleaving is performed with respect to the error-correction-coded block, the locations of the bits in the error-correction-coded block may be changed.

FIG. 23 is a diagram showing the concept of another embodiment of interleaving of a bit interleaver. In the embodiment shown in this drawing, when bits are written in a memory, the bits are written in a column direction. When the written bits are read, the bits of the circularly shifted locations are read in a row direction. In each row, the bits written in each row is circularly shifted. If the bits are written or read by a circular shift method with respect to the row or the column of the memory, this is called twisted bit interleaving. This embodiment relates to the twisted bit interleaving method using a method of reading the bits after the bits are shifted by one column in row direction. Instead of shifting the written bits in the memory, the point for reading bits in the memory or the point for writing bits in the memory can be shifted.

In this embodiment, N denotes the length of the error correction coded block and C denotes the length of the column. When the bits are written, the bits are written in a first column (represented by a shadow) in order of 1, 2, 3, 4, . . . , and C and the bits are written in a second column in order of C+1, C+2, C+3, . . . .

The written bits are twisted in the row direction one column by one column.

If the written bits are read, the twisted bits are read in the row direction. For example, in this embodiment, the bits are read in a first row in order of 1, C+1, . . . and the bits are read in a second row in order of X1, 2, C+2, . . . (X1 is a bit in the first column of the second row). The bits are read by row by row and the circularly shifted bits are read. Of course, instead of shifting the written bits in the memory, the point for reading bits written in the memory can be shifted.

FIG. 24 is a view showing another embodiment of bit interleaving. In this embodiment, N denotes the length of the error correction coded block and C denotes the length of the column. When the bits are written, the bits are written in a first column in order of 1, 2, 3, 4, . . . , C-1, and C and the bits are written in a second column in order of C+1, C+2, C+3, . . . .

The written bits are double-twisted in the row direction two columns by two columns. If the written bits are read, the bits circularly shifted by two columns are read in the column direction in every row. This method may be called a double twisted bit interleaving method.

FIG. 25 is a view showing another embodiment of bit interleaving. In this embodiment, N denotes the length of the error correction coded block and C denotes the length of the column. The bits are written in a first column in order of 1, 2, 3, 4, . . ., C-1, and C and the bits are written in a second column in order of C+1, C+2, C+3, . . . .

When the written bits are read, in a first region of the rows, the bits may be read by the twisted bit interleaving method.

In a second region of the rows, the bits may be read by the double twisted interleaving method.

In a third region of the rows, the bits may be read by the twisted bit interleaving method.

If the bits are interleaved by at least one of the twisted bit interleaving method and the double twisted interleaving method, the bits in the error correction coded block can be more randomly mixed.

FIG. 26 is a view showing the concept of multiplexing of input bits of the demuxs 1313 a and 1313 b.

The bit interleavers 1312 a and 1312 b interleave the input bits x0, x1, . . . , and xn−1 and output the interleaved bits. The interleaving method is already described above.

The demuxs 1313 a and 1313 b demultiplex the interleaved bit streams. The demultiplexing method may vary according to the code rate of the error correction coding method and the symbol mapping method of the symbol mapper. If the symbol method of the symbol mapper is QPSK, the input bits, for example, are interleaved to two sub streams and the symbol mapper maps the two sub streams to the symbols so as to correspond to the real axis and the imaginary axis of the constellation. For example, a first bit y0 of the demultiplexed first sub stream corresponds to the real axis and a first bit yl of the demultiplexed second sub stream corresponds to the imaginary axis.

If the symbol method of the symbol mapper is 16 QAM, the input bits, for example, are demultiplexed to four sub frames. The symbol mapper selects the bits included in the four sub streams and maps the selected bits to the symbols so as to correspond to the real axis and the imaginary axis of the constellation.

For example, the bits y0 and y2 of the demultiplexed first and third sub streams correspond to the real axis and the bits y1 and y3 of the demultiplexed second and fourth sub streams correspond to the imaginary axis.

Similarly, if the symbol method of the symbol mapper is 64 QAM, the input bits may be demultiplexed to six bit streams. The symbol mapper maps the six sub streams to the symbols so as to correspond to the real axis and the imaginary axis of the constellation. For example, the demultiplexed first, third and fifth sub stream bits y0, y2 and y4 correspond to the real axis and the demultiplexed second, fourth and sixth sub stream bits y1, y3 and y6 correspond to the imaginary axis.

Similarly, if the symbol method of the symbol mapper is 256 QAM, the input bits may be demultiplexed to eight bit streams. The symbol mapper maps the eight sub streams to the symbols so as to correspond to the real axis and the imaginary axis of the constellation. For example, first, the demultiplexed first, third fifth and seventh sub stream bits y0, y2, y4 and y6 correspond to the real axis and the demultiplexed second, fourth, sixth and eighth sub stream bits y1, y3, y6 and y7 correspond to the imaginary axis.

If the symbol mapper maps the symbols, the sub streams demultiplexed by the demux may be mapped to the bit streams of the real axis and the imaginary axis of the constellation.

The above-described bit interleaving method, demultiplexing method and symbol mapping method are exemplary and various methods may be used as the method of selecting the bits in the sub streams such that the sub streams demultiplexed by the demux may correspond to the real axis and the imaginary axis of the constellation.

The cell word mapped to the symbols may vary according to any one of the error-corrected bit streams according to the code rate, the method of interleaving the bit streams, the demultiplexing method and the symbol mapping method. The MSB of the cell word is higher than the LSB of the cell word in the reliability of the error correction decoding. Although the reliability of the bit of a specific location of the error-correction-coded block is low, the reliability of the bit can be improved by the symbol demapping process if the bit of the cell word is arranged on the MSB or close to the MSB.

Accordingly, although the reliability of the bit coded according to the characteristics of the H-matrix used in the irregular LDPC error correction coding method is changed, the bit can be robustly transmitted/received by the symbol mapping and demapping process and the system performance can be adjusted.

FIG. 27 is a view showing an embodiment of demultiplexing an input stream by the demux.

If the symbol mapping method is QPSK, two bits are mapped to one symbol and the two bits of one symbol unit are demultiplexed in order of the bit indexes (indexes 0 and 1 of b).

If the symbol mapping method is 16 QAM, 4 bits are mapped to one symbol and the four bits of one symbol unit are demultiplexed according to the calculating result of the modulo-4 of bit indexes (indexes 0, 1, 2 and 3 of b).

If the symbol mapping method is 64 QAM, 6 bits are mapped to one symbol and the six bits of one symbol unit are demultiplexed according to the calculating result of the modulo-6 of bit indexes (indexes 0, 1, 2, 3, 4 and 5 of b).

If the symbol mapping method is 256 QAM, 8 bits are mapped to one symbol and the eight bits of one symbol unit are demultiplexed according to the calculating result of the modulo-8 of bit indexes (indexes 0, 1, 2, 3, 4, 5, 6 and 7 of b).

The demultiplexing order of the sub streams is exemplary and may be modified.

FIG. 28 is a view showing an example of a demultiplexing type according to a symbol mapping method. The symbol mapping method includes QPSK, 16 QAM, 64 QAM and 256 QAM, and the demultiplexing type includes a first type to a sixth type.

The first type is an example in which the input bits sequentially correspond to even-numbered indexes (0, 2, 4, 8, . . . ) (or the real axis of the constellation) and sequentially correspond to odd-numbered indexes (1, 3, 5, 7, . . . ) (or the imaginary axis of the constellation). Hereinafter, the bit demultiplexing of the first type may be represented by a demultiplexing identifier 10 (a binary number of 1010; the location of 1 is the location of the MSB corresponding to the real axis and the imaginary axis of the constellation).

The second type is an example in which the demultiplexing is performed in reverse order of the first type, that is, the LSB of the input bits sequentially correspond to even-numbered indexes (6, 4, 2, 0) (or the real axis of the constellation) and odd-numbered indexes (1, 3, 5, 7, . . . ) (or the imaginary axis of the constellation). Hereinafter, the bit demultiplexing of the second type may be represented by a demultiplexing identifier 5 (a binary number of 0101).

The third type is an example in which the input bits are arranged such that the bits of the both ends of the codeword become the MSB. The input bits are rearranged so as to fill the code word from the both ends of the code word. Hereinafter, the bit demultiplexing of the third type may be represented by a demultiplexing identifier 9 (a binary number of 1001).

The fourth type is an example in which the input bits are arranged such that a middle bit of the code word becomes the MSB. A bit of the input bits is first filled in the middle location of the code word and the remaining bits are then rearranged toward the both ends of the code word in order of the input bits. Hereinafter, the bit demultiplexing of the fourth type may be represented by a demultiplexing identifier 6 (a binary number of 0110).

The fifth type is an example in which the bits are demultiplexed such that a last bit of the code word becomes the MSB and a first bit thereof becomes the LSB, and the sixth type is an example in which the bits are rearranged such that the first bit of the code word becomes the MSB and the last bit thereof becomes the LSB. Hereinafter, the bit demultiplexing of the fifth type may be represented by a demultiplexing identifier 3 (a binary number of 0011), and the bit demultiplexing of the sixth type may be represented by a demultiplexing identifier 12 (a binary number of 1100).

As described above, the demultiplexing type may vary according to the symbol mapping method or the code rate of the error correction coding method. That is, a different demultiplexing type may be used if the symbol mapping method or the code rate is changed.

FIG. 29 is a view showing an embodiment of demultiplexing an input bit stream according to a demultiplexing type. This embodiment may include bit interleavers 1312 a and 1312 b, demuxs 1313 a and 1313 b and mappers 1315 a and 1315 b.

The bit interleavers 1312 a and 1312 b interleave the error-correction-coded PLP service streams. For example, the bit interleavers 1312 a and 1312 b may perform the bit interleaving in the error correction coding units according to the error correction coding mode. The bit interleaving method is already described above.

The demuxs 1313 a and 1313 b may include first type demuxs 1313 a 1 and 1313 b 1, . . . , and n^(th) type demuxs 1313 a 2 and 1313 b 2. Here, n is an integer. The methods of demultiplexing the bits by the n types of demuxs follow the types shown in FIG. 17. For example, the first type demuxs may correspond to the first type bit demultiplexing (1100) and the second type demux (not shown) may correspond to the second type bit demultiplexing (0011). The n^(th) type demux 1313 b demultiplexes the input bit stream according to the n^(th) type bit multiplexing (e.g., the demultiplexing identifier 1100) and outputs the demultiplexed bit stream. Selectors 1313 a 3 and 1313 b 3 receive a demux selection signal of the demultiplexing type suitable for the input bits and output the demultiplexed bit stream according to any one of the first type to the n^(th) type and the demux selection signal. The demux selection signal may vary according to the code rate of the error correction coding and the symbol mapping method of the constellation. Accordingly, the demultiplexing type may be determined according to the code rate of the error correction coding method or/and the symbol mapping method of the constellation. The detailed example according to the symbols mapped to the constellation or/and the code rate of the error correction coding according to the demux selection signal will be described later.

The mappers 1315 a and 1315 b may map the demultiplexed sub bit streams to the symbols according to the demux selection signal and output the mapped symbols.

FIG. 30 is a view showing a demultiplexing type which is determined according to a code rate of the error correction coding and the symbol mapping method.

In the 4 QAM symbol mapping method, even when the code rate cr of the LDPC error correction coding method is any one of 1/4, 1/3, 2/5, 1/2, 3/5, 2/3, 3/4, 4/5, 5/6, 8/9 and 9/10, the bit stream can be demultiplexed according to all the demultiplexing types (denoted by all).

In the 16 QAM symbol mapping method, if the code rate of the LDPC error correction coding method is 1/4, 1/3, 2/5 and 1/2, the symbols can be mapped without performing the bit interleaving and the bit demultiplexing (denoted by No-Int and No-Demux). If the code rate of the error correction coding is 3/5, the bit can be demultiplexed according to any one of the demultiplexing identifiers 9, 10 and 12. If the code rate of the error correction coding is 2/3, 3/4, 4/5, 5/6, 8/9 and 9/10, the input bit stream can be demultiplexed according to the demultiplexing identifier 6.

In the 64 QAM symbol mapping method, if the code rate of the LDPC error correction coding is 1/4, 1/3, 2/5 and 1/2, the symbols can be mapped without performing the bit interleaving and the bit demultiplexing. If the code rate is 3/5, the bits can be demultiplexed according to any one of the demultiplexing identifiers 9 and 10. If the code rate is 2/3, 3/4, 4/5, 5/6, 8/9 and 9/10, the bits can be demultiplexed according to the demultiplexing identifier 6.

In the 256 QAM symbol mapping method, if the code rate of the LDPC error correction coding is 1/4, 1/3, 2/5 and 1/2, the symbols can be mapped without performing the bit interleaving and the bit demultiplexing. If the code rate is 3/5, the bits can be demultiplexed according to the demultiplexing identifier 9. If the code rate is 2/3, 3/4, 4/5, 5/6, 8/9 and 9/10, the bits can be demultiplexed according to the demultiplexing identifier 6.

As described above, the bit demultiplexing type may vary according to the code rate used for the error correction coding and the symbol mapping method. Accordingly, the error correction capability of a bit located on a specific location of the error-correction-coded block may be adjusted by mapping the demultiplexed sub streams to the symbols. Accordingly it is possible to optimize the robustness in the bit level.

FIG. 31 is a view showing an example of expressing the demultiplexing method by an equation. For example, if the symbol mapping method is QPSK, the input bits (x_(i), x_(N/2+i)) correspond to the demultiplexed bits y0 and y1. If the symbol mapping method is 16 QAM, the input bits

$\left( {x_{\frac{2\; N}{4} + i},x_{\frac{3\; N}{4} + i},x_{i},x_{\frac{n}{4} + i}} \right)$ correspond to the demultiplexed bits y0, y1, y2 and y3.

If the symbol mapping method is 64 QAM, the input bits

$\left( {x_{\frac{4\; N}{6} + i},x_{\frac{5\; N}{6} + i},x_{\frac{2N}{6} + i},x_{\frac{3N}{6} + i},x_{i},x_{\frac{N}{6} + i}} \right)$ correspond to the demultiplexed bits y0, y1, y2, y3, y4 and y5. If the symbol mapping method is 256 QAM, the input bits

$\left( {x_{\frac{6\; N}{8} + i},x_{\frac{7\; N}{8} + i},x_{\frac{4N}{8} + i},x_{\frac{5N}{8} + i},x_{\frac{2N}{8} + i},x_{\frac{3N}{8} + i},x_{i},x_{\frac{N}{8} + i}} \right)$ correspond to the demultiplexed bits y0, y1, y2, y3, y4, y5, y6 and y7.

Here, N denotes the number of bits mapped to the symbols with respect to the input of the bit interleaver.

FIG. 32 is a view showing an example of mapping a symbol by a symbol mapper. For example, in the QPSK symbol mapping method, the symbols on the constellation correspond to the value of the bit y0 of the demultiplexed first sub stream and the value of the bit y1 of the demultiplexed second sub stream.

In the 16 QAM, the real axis of the symbols on the constellation corresponds to the bits of the demultiplexed first and third sub streams (bits separated from the location of the MSB by 0 and 2) and the imaginary axis thereof corresponds to the bits of the demultiplexed second and fourth sub streams (bits separated from the location of the MSB by 1 and 3).

In the 64 QAM, the real axis of the symbols on the constellation corresponds to the bits of the demultiplexed first, third, and fifth sub streams (bits separated from the location of the MSB by 0, 2 and 4) and the imaginary axis thereof corresponds to the bits of the demultiplexed second, fourth and sixth sub streams (bits separated from the location of the MSB by 1, 3 and 5).

Accordingly, the bits configuring the symbol may be mapped to the cell word in the demultiplexing order. If the bits configuring the cell word are demultiplexed, the MSB and the LSB of the cell word are changed and the robustness of the bits can be adjusted although the reliabilities of the LDPC error-correction-coded bits vary according to the locations.

FIG. 33 is a block diagram illustrating a MIMO/MISO encoder according to an embodiment of the present invention. The MIMO/MISO encoder encodes the input data using the MIMO/MISO encoding scheme, and outputs the encoded data to several paths. If a signal reception end receives the signal transmitted to the several paths from one or more paths, it is able to acquire a gain (also called a diversity gain, a payload gain, or a multiplexing gain).

The MIMO/MISO encoder 140 encodes service data of each path generated from the frame builder 130, and outputs the encoded data to the A number of paths corresponding to the number of output antennas.

FIG. 34 is a block diagram illustrating a modulator according to an embodiment of the present invention. The modulator includes a first power controller (PAPR Reduce1) 151, a time-domain transform unit (IFFT) 153, a second power controller (PAPR Reduce2) 157, and a guard-interval inserter 159.

The first power controller 151 reduces a PAPR (Peak-to-Average Power Ratio) of data transmitted to the R number of signal paths in the frequency domain.

The time-domain transform (IFFT) unit 153 converts the received frequency-domain signals into time-domain signals. For example, the frequency-domain signals may be converted into the time-domain signals according to the IFFT algorithm. Therefore, the frequency-domain data may be modulated according to the OFDM scheme.

The second power controller (PAPR Reduce2) 157 reduces a PAPR (Peak-to-Average Power Ratio) of channel data transmitted to the R number of signal paths in the time domain. In this case, a tone reservation scheme, and an active constellation extension (ACE) scheme for extending symbol constellation can be used.

The guard-interval inserter 159 inserts the guard interval into the output OFDM symbol, and outputs the inserted result. As described above, the above-mentioned embodiment can be carried out in each signal of the R number of paths.

FIG. 35 is a block diagram illustrating an analog processor 160 according to an embodiment of the present invention. The analog processor 160 includes a digital-to-analog converter (DAC) 161, an up-conversion unit 163, and an analog filter 165.

The DAC 161 converts the input data into an analog signal, and outputs the analog signal. The up-conversion unit 163 converts a frequency domain of the analog signal into an RF area. The analog filter 165 filters the RF-area signal, and outputs the filtered RF signal.

FIG. 36 is a block diagram illustrating an apparatus for receiving a signal according to an embodiment of the present invention. The signal reception apparatus includes a first signal receiver 210 a, an n-th signal receiver 210 n, a first demodulator 220 a, an n-th demodulator 220 n, a MIMO/MISO decoder 230, a frame parser 240, and a decoding demodulator 250, and an output processor 260.

In the case of a reception signal according to the TFS signal frame structure, several services are multiplexed to R channels, and are then time-shifted, such that the time-shifted result is transmitted.

The receiver may include at least one signal receiver for receiving a service transmitted over at least one RF channel. The TFS signal frame transmitted to the R (where R is a natural number) number of RF channels can be transmitted to a multi-path via the A number of antennas. The A antennas have been used for the R RF channels, such that a total number of antennas is R×A.

The first signal receiver 210 a is able to receive service data transmitted via at least one path from among overall service data transmitted via several RF channels. For example, the first signal receiver 210 a can receive the transmission signal processed by the MIMO/MISO scheme via several paths.

The first signal receiver 210 a and the n-th signal receiver 210 n can receive several service data units transmitted over n number of RF channels from among several RF channels, as a single PLP. Namely, this embodiment shows the signal reception apparatus capable of simultaneously receiving data of the R number of RF channels. Therefore, if this embodiment receives a single RF channel, only the first receiver 210 a is needed.

The first demodulator 220 a and the n-th demodulator 220 n demodulate signals received in the first and n-th signal receivers 210 a and 210 n according to the OFDM scheme, and output the demodulated signals.

The MIMO/MISO decoder 230 decodes service data received via several transmission paths according to the MIMO/MISO decoding scheme, and outputs the decoded service data to a single transmission path. If the number R of services transmitted over several transmission paths are received, the MIMO/MISO decoder 230 can output single PLP service data contained in each of R services corresponding to the number of R channels. If P number of services are transmitted via the R number of RF channels, and signals of individual RF channels are received via the A number of antennas, the receiver decodes the P number of services using a total of (R×A) reception antennas.

The frame parser 240 parses the TFS signal frame including several services, and outputs the parsed service data.

The decoding demodulator 250 performs the error correction decoding on the service data contained in the parsed frame, demaps the decoded symbol data into bit data, and outputs the demapping-processed result.

The output processor 260 decodes a stream including the demapped bit data, and outputs the decoded stream.

In the above-mentioned description, each of the frame parser 240, and the decoding demodulator 250, and the output processor 260 receives several service data units as many as the number of PLPs, and performs signal processing on the received service data.

FIG. 37 is a block diagram illustrating a signal receiver according to an embodiment of the present invention. The signal receiver may include a tuner 211, a down-converter 213, and an analog-to-digital converter (ADC) 215.

The tuner 211 performs hopping of some RF channels capable of transmitting user-selected services in all RF channels when the PLP is included in several RF channels, and outputs the hopping result. The tuner 211 performs hopping of RF channels contained in the TFS signal frame according to input RF center frequencies, and at the same time tunes corresponding frequency signals, such that it outputs the tuned signals. If a signal is transmitted to A number of multi-paths, the tuner 211 performs the tuning to a corresponding RF channel, and receives reception signals via the A number of antennas.

The down converter 213 performs down conversion of the RF frequency of the signal tuned by the tuner 211, and outputs the down-conversion result. The ADC 215 converts an analog signal into a digital signal.

FIG. 38 is a block diagram illustrating a demodulator according to an embodiment of the present invention. The demodulator includes a frame detector 221, a frame synchronization unit 222, a guard-interval remover 223, a frequency-domain transform unit (FFT) 224, a channel estimator 225, a channel equalizer 226, and a signaling-information extractor 227.

If the demodulator acquires service data transmitted to a single PLP stream, the following signal demodulation will be carried out. A detailed description thereof will hereinafter be described.

The frame detector 221 identifies a delivery system of a reception signal. For example, the frame detector 221 determines whether the reception signal is a DVB-TS signal or not. And, the frame detector 221 may also determine whether a reception signal is a TFS signal frame or not. The frame synchronization unit 222 acquires time- and frequency-domain synchronization of the TFS signal frame.

The guide interval controller 223 removes a guard interval located between OFDM symbols from the time domain. The frequency-domain converter (FFT) 224 converts a reception signal into a frequency-domain signal using the FFT algorithm, such that it acquires frequency-domain symbol data.

The channel estimator 225 performs channel estimation of a reception channel using a pilot symbol contained in symbol data of the frequency domain. The channel equalizer 226 performs channel equalization of reception data using channel information estimated by the channel estimator 225.

The signaling information extractor 227 can extract the signaling information of a physical layer established in the first and second pilot signals contained in channel-equalized reception data.

FIG. 39 is a block diagram illustrating a MIMO/MISO decoder according to an embodiment of the present invention. The signal receiver and the demodulator are designed to rocess a signal received in a single path. If the signal receiver and the demodulator receive PLP service data providing a single service via several paths of several antennas, and demodulate the PLP service data, the MIMO/MIMO decoder 230 outputs the signal received in several paths as service data transmitted to a single PLP. Therefore, the MIMO/MISO decoder 230 can acquire a diversity gain and a multiplexing gain from service data received in a corresponding PLP.

The MIMO/MISO decoder 230 receives a multi-path transmission signal from several antennas, and is able to decode a signal using a MIMO scheme capable of recovering each reception signal in the form of a single signal. Otherwise, the MIMO/MISO decoder 230 is able to recover a signal using a MIMO scheme which receives the multi-path transmission signal from a single antenna and recovers the received multi-path transmission signal.

Therefore, if the signal is transmitted via the R number of RF channels (where R is a natural number), the MIMO/MISO decoder 230 can decode signals received via the A number of antennas of individual RF channels. If the A value is equal to “1”, the signals can be decoded by the MISO scheme. If the A value is higher than “1”, the signals can be decoded by the MIMO scheme.

FIG. 40 is a block diagram illustrating a frame parser according to an embodiment of the present invention. The frame parser includes a first frequency de-interleaver 241 a, a r-th frequency de-interleaver 241 r, a frame parser 243, a first time de-interleaver 245 a, a p-th time de-interleaver 245 p, a first symbol demapper 247 a, and a p-th symbol demapper. The value of “r” can be decided by the number of RF channels, and the value of “p” can be decided by the number of streams transmitting PLP service data generated from the frame parser 243.

Therefore, if p number of services are transmitted to p number of PLP streams over R number of RF channels, the frame parser includes the r number of frequency de-interleavers, the p number of time de-interleavers, and the p number of symbol demappers.

In association with a first RF channel, the first frequency interleaver 241 a performs de-interleaving of frequency-domain input data, and outputs the de-interleaving result.

The frame parser 243 parses the TFS signal frame transmitted to several RF channels using scheduling information of the TFS signal frame, and parses PLP service data contained in the slot of a specific RF channel including a desired service. The frame parser 243 parses the TFS signal frame to receive specific service data distributed to several RF channels according to the TFS signal frame structure, and outputs first-path PLP service data.

The first time de-interleaver 245 a performs de-interleaving of the parsed first-path PLP service data in the time domain. The first symbol demapper 247 a determines service data mapped to the symbol to be bit data, such that it can output a PLP stream associated with the first-path PLP service data.

Provided that symbol data is converted into bit data, and each symbol data includes symbols based on the hybrid symbol-mapping scheme, the p number of symbol demappers, each of which includes the first symbol demapper, can determine the symbol data to be bit data using different symbol-demapping schemes in individual intervals of the input symbol data.

FIG. 41 is a view showing an embodiment of each of symbol demappers 247 a and 247 p. The symbol demappers receive the streams corresponding to the PLPs from the time interleavers 245 a and 245 p respectively corresponding to the symbol demappers.

Each of the symbol demappers 247 a and 247 p may include an error correction block splitter 2471, a symbol splitter 2473, a first order demapper 2475 a, a second order demapper 2475 b and a bit stream merger 2478.

The error correction block splitter 2471 may split the PLP stream received from the corresponding one of the time interleavers 245 a and 245 p in the error correction block units. The error correction block splitter 2471 may split the service stream in the normal mode LDPC block unit. In this case, the service stream may be split in a state in which four blocks according to the short mode (the block having the length of 16200 bits) are treated as the error correction block of one block according to the normal mode (the block having the length of 64800 bits).

The symbol splitter 2473 may split the symbol stream in the split error correction block according to the symbol mapping method of the symbol stream.

For example, the first order demapper 2475 a converts the symbols according to the higher order symbol mapping method into the bits. The second order demapper 2475 b converts the symbols according to the lower order symbol mapping method into the bits.

The bit stream merger 2478 may receive the converted bits and output one bit stream.

FIG. 42 is a view showing another embodiment of each of the symbol demappers 247 a and 247 p. The embodiment of this drawing is similar to the embodiment of FIG. 41 except that a first order power calibration unit 2474 a and a second order power calibration unit 2474 b are further included.

The first order power calibration unit 2474 a receives the symbols split by the symbol splitter 2473, calibrates the power of the received symbols according to the symbol mapping schemes, and outputs the calibrated symbols. The power of the received symbols may have the power calibrated according to the size of the constellation based on the symbol mapping methods. The first order power calibration unit 2474 a converts the power calibrated in accordance with the into the original symbol power of the constellation. The first order demapper 2475 a may demap the symbols, of which the power is calibrated by the first order power calibration unit, to the bits.

Similarly, the second order power calibration unit 2474 b receives the symbols split by the symbol splitter 2473, modified the calibrated power of the received symbols to the original power according to the size of the constellation, and outputs the modified symbols.

FIG. 43 is a view showing another embodiment of each of the symbol demappers 247 a and 247 p. Each of the symbol demappers 247 a and 247 p may include a symbol splitter 2473, a first order demapper 2474 a, a second order demapper 2474 b, a first order mux 2475 a, a second order mux 2475 b, a first order bit deinterleaver 2476 a, a second order bit deinterleaver 2476 b and a bit stream merger 2478. By this embodiment, the embodiment of the decoding and demodulation unit of FIG. 33 includes a first decoder 253, a first deinterleaver 255 and a second decoder 257.

The symbol splitter 2473 may split the symbol stream of the PLP according to the method corresponding to the symbol mapping method.

The first order demapper 2474 a and the second order demapper 2474 b convert the split symbol streams into bits. For example, the first order demapper 2474 a performs the symbol demapping of the higher order QAM and the second order demapper 2474 b performs the symbol demapping of the lower order QAM. For example, the first order demapper 2474 a may perform the symbol demapping of 256 QAM and the second order demapper 2474 b may perform the symbol demapping of 64 QAM.

The first order mux 2475 a and the second order mux 2475 b multiplex the symbol-mapped bits. The multiplexing methods may correspond to the demultiplexing methods described with reference to FIGS. 15 to 18. Accordingly, the demultiplexed sub streams may be converted into one bit stream.

The first order bit deinterleaver 2476 a deinterleaves the bit streams multiplexed by the first order mux 2475 a. The second order bit deinterleaver 2476 b deinterleaves the bits multiplexed by the first order mux 2475 a. The deinterleaving method corresponds to the bit interleaving method. The bit interleaving method is shown in FIG. 12.

The bit stream merger 2478 may merge the bit streams deinterleaved by the bit interleavers 2476 a and 2476 b to one bit stream.

The first decoder 253 of the decoding and demodulation unit may error correction decode the output bit stream according to the normal mode or the short mode and the code rate according to the modes.

FIG. 44 is a view showing another embodiment of each of the symbol demappers 247 a and 247 p. The embodiment of this drawing is similar to the embodiment of FIG. 43 except that a first order power calibration unit 2474 a and a second order power calibration unit 2474 b are further included. The first order power calibration unit 2474 a and the second order power calibration unit 2474 b modify the calibrated powers of the symbols according to the symbol mapping methods and output the modified symbols to the symbol demappers 2475 a and 2475 b.

FIG. 45 is a view showing an embodiment of multiplexing the demultiplexed sub stream. In this embodiment, the demappers 2474 a and 2474 b decide the cell words including the bits. The muxs 2475 a and 2475 b multiplex the decided cell words according to the mux selection signal. The demultiplexed cell words are input to any one of first muxs 2475 a 2 and 2475 b 2 to n^(th) muxs 2475 a 3 and 2475 b 3.

The first muxs 2475 a 2 and 2475 b 2 to the n^(th) muxs 2475 a 3 and 2475 b 3 change the order of the bits in the cell words input according to the mux selection signal. The mux selection signal may be changed according to the code rate of the error correction coding or the symbol mapping method. In order to generate one stream and the bit streams delivered to the muxs, the order of selecting the sub stream may be changed according to the mux selection signal.

The first demuxs 2475 a 1 and 2475 b 1 output the symbol-demapped bit streams to any one of the first muxs 2475 a 2 and 2475 b 2 to the n^(th) muxs 2475 a 3 and 2475 b 3 according to the mux selection signal. The first sub muxs 2475 a 1 and 2475 b 1 may receive the sub streams multiplexed by the first muxs 2475 a 2 and 2475 b 2 to the n^(th) muxs 2475 a 3 and 2475 b 3 and output one stream, according to the mux selection signal.

The cell words including the changed bits are input to the bit interleavers 2476 a and 2476 b, and the bit deinterleavers 2476 a and 2476 b deinterleave the input bits and output the deinterleaved bits.

FIG. 46 is a block diagram illustrating a decoding demodulator according to an embodiment of the present invention. The decoding demodulator may include several function blocks corresponding to the coding and modulation unit. In this embodiment, the decoding demodulator of FIG. 16 may include a first de-interleaver 251, a first decoder 253, a second de-interleaver 255, and a second decoder 257. The second de-interleaver 255 can be selectively contained in the decoding demodulator.

The first de-interleaver 251 acts as an inner de-interleaver, and is able to perform de-interleaving of the p-th PLP stream generated from the frame parser.

The first decoder 253 acts as an inner decoder, can perform error correction of the de-interleaved data, and can use an error correction decoding algorithm based on the LDPC scheme.

The second de-interleaver 255 acts as an outer interleaver, and can perform de-interleaving of the error-correction-decoded data.

The second decoder 257 acts as an outer decoder. Data de-interleaved by the second de-interleaver 255 or error-corrected by the first decoder 253 is error-corrected again, such that the second decoder 257 outputs the re- error-corrected data. The second decoder 257 decodes data using the error correction decoding algorithm based on the BCH scheme, such that it outputs the decoded data.

The first de-interleaver 251 and the second de-interleaver 255 are able to convert the burst error generated in data contained in the PLP stream into a random error. The first decoder 253 and the second decoder 257 can correct errors contained in data.

The decoding demodulator shows operation processes associated with a single PLP stream. If the p number of streams exist, the p number of decoding demodulators are needed, or the decoding demodulator may repeatedly decode input data p times.

FIG. 47 is a block diagram illustrating an output processor according to an embodiment of the present invention. The output processor may include p number of baseband (BB) frame parsers (251 a, . . . , 261 p), a first service merger 263 a, a second service merger 263 b, a first demultiplexer 265 a, and a second demultiplexer 265 b.

The BB frame parsers (261 a, . . . , 261 p) remove BB frame headers from the first to p-th PLP streams according to the received PLP paths, and output the removed result. This embodiment shows that service data is transmitted to at least two streams. A first stream is an MPEG-2 TS stream, and a second stream is a GS stream.

The first service merger 263 a calculates the sum of service data contained in payload of at least one BB frame, such that it outputs the sum of service data as a single service stream. The first demultiplexer 255 a may demultiplex the service stream, and output the demultiplexed result.

In this way, the second service merger 263 b calculates the sum of service data contained in payload of at least one BB frame, such that it can output another service stream. The second demultiplexer 255 b may demultiplex the GS-format service stream, and output the demultiplexed service stream.

FIG. 48 is a block diagram illustrating an apparatus for transmitting a signal according to an embodiment of another embodiment of an embodiment of by the present invention. The signal transmission apparatus includes a service composer 310, a frequency splitter 320, and a transmitter 400. The transmitter 400 encodes or modulates a signal including a service stream to be transmitted to each RF band.

The service composer 310 receives several service streams, multiplexes several service streams to be transmitted to individual RF channels, and outputs the multiplexed service streams. The service composer 310 outputs scheduling information, such that it controls the transmitter 400 using the scheduling information, when the transmitter 400 transmits the PLP via several RF channels. By this scheduling information, the service composer 310 modulates several service frames to be transmitted to the several RF channels by the transmitter 400, and transmits the modulated service frames.

The frequency splitter 320 receives a service stream to be transmitted to each RF band, and splits each service stream into several sub-streams, such that the individual RF frequency bands can be allocated to the sub-streams.

The transmitter 400 processes the service streams to be transmitted to individual frequency bands, and outputs the processed resultant streams. For example, in association with a specific service stream to be transmitted to the first RF channel, the first mapper 410 maps the input service stream data into symbols. The first interleaver 420 interleaves the mapped symbols to prevent the burst error.

The first symbol inserter 430 can insert a signal frame equipped with a pilot signal (e.g., a scatter pilot signal or a continual pilot signal) into the modulated signal.

The first modulator 440 modulates the data interleaved by the signal modulation scheme. For example, the first modulator 440 can modulate signals using the OFDM scheme.

The first pilot symbol inserter 450 inserts the first pilot signal and the second pilot signal in the signal frame, and is able to transmit the TFS signal frame.

Service stream data transmitted to the second RF channel is transmitted to the TFS signal frame via several blocks 415, 425, 435, 445, and 455 of different paths shown in the transmitter of FIG. 18.

The number of signal processing paths transmitted from the transmitter 400 may be equal to the number of RF channels contained in the TFS signal frame.

The first mapper 410 and the second mapper may respectively include the demultiplexers 1313 a and 1313 b, and allow the locations of the MSB and the LSB to be changed in the symbol-mapped cell word.

FIG. 49 is a block diagram illustrating an apparatus for receiving a signal according to another embodiment of the present invention. The signal reception apparatus may include a reception unit 510, a synchronization unit 520, a mode detector 530, an equalizer 540, a parameter detector 550, a de-interleaver 560, a demapper 570, and a service decoder 580.

The reception unit 500 is able to receive signals of a first RF channel selected by a user from among the signal frame. If the signal frame includes several RF channels, the reception unit 500 performs hopping of the several RF channels, and at the same time can receive a signal including the selected service frame.

The synchronization unit 510 acquires synchronization of a reception signal, and outputs the synchronized reception signal. The demodulator 520 is able to demodulate the synchronization-acquired signal. The mode detector 530 can acquire a FFT mode (e.g., 2k, 4k, 8k FFT operation length) of the second pilot signal using the first pilot signal of the signal frame.

The demodulator 520 demodulates the reception signal under the FFT mode of the second pilot signal. The equalizer 540 performs channel estimation of the reception signal, and outputs the channel-estimation resultant signal. The de-interleaver 560 de-interleaves the channel-equalized reception signal. The demapper 570 demaps the interleaved symbol using the symbol demapping scheme corresponding to the transmission-signal symbol mapping scheme (e.g., QAM).

The parameter detector 550 acquires physical parameter information (e.g., Layer-1 (L1) information) contained in the second pilot signal from the output signal of the equalizer 540, and transmits the acquired physical parameter information to the reception unit 500 and the synchronization unit 510. The reception unit 500 is able to change the RF channel to another channel using network information detected by the parameter detector 550.

The parameter detector 550 outputs service-associated information, service decider 580 decodes service data of the reception signal according to the service-associated information from the parameter detector 550, and outputs the decoded service data.

The demapper 570 may include the muxs 2475 a and 2475 b and output the bit stream obtained by restoring the order of the bits of which the locations of the MSB and the LSB are changed according to the code rate of the error correction coding and the symbol mapping method.

Hereinafter, a method for modulating a first pilot signal of a signal frame having at least one RF band and a method and apparatus for receiving the modulated first pilot signal will be described.

The time-interleaved PLP symbols are transmitted via regions, which are temporally divided in the signal frame. The time-interleaved PLP symbols may be transmitted via regions, which are divided in the frequency domain, if a plurality of RF bands exists. Accordingly, if the PLP is transmitted or received, a diversity gain can be obtained. An error correction mode and a symbol mapping method may be changed according to services corresponding to transport streams or may be changed in the service.

A first pilot signal and a second pilot signal are arranged at the start location of the signal frame having such characteristics, as a preamble signal.

As described above, the first pilot signal included in the signal frame may include an identifier for identifying the signal frame having the above-described structure. The first pilot signal may include information about the transmission structure indicating whether or not the signal frame is transmitted via multiple paths and information about an FFT mode of a signal following the first pilot signal. The receiver can detect the signal frame from the first pilot signal and obtain the information about the integral carrier frequency offset estimation and information about the FFT mode of the data symbol.

FIG. 50 is a view showing an embodiment of the structure of a first pilot signal. A portion denoted by A is a useful portion of the first pilot signal. B denotes the same cyclic prefix as a first portion of the portion A in the time domain and C denotes the same cyclic suffix as a second portion of the portion A in the time region. The first portion may be duplicated from the second half of the portion A and the second portion may be duplicated from the first half of the portion A.

B and C can be respectively obtained by duplicating the first portion and the second portion and frequency shifting the duplicated portions. A relationship between B or C and A is as follows. B=onepart(A)·e ^(j2πf) ^(SH) ^(t)  [Equation 1] C=anotherpart(A)·e ^(j2πf) ^(SH) ^(t)

In the above equation, SH denotes a shift unit of the frequency shift. Accordingly, the frequency shift values of the portions B and C may be inversely proportional to the lengths of the portions B and C.

If the first pilot signal is configured by frequency shifting the cyclic prefix (B) and the cyclic suffix (C), the probability that the data symbol is erroneously detected to the preamble is low and the probability that the preamble is erroneously detected is reduced, although the data symbols configuring the PLP and the symbols configuring the preamble are modulated in the same FFT mode.

If continuous wave (CW) interference is included like an analog TV signal, the probability that the preamble is erroneously detected due to a noise DC component generated in a correlation process, is reduced. In addition, if the size of the FFT applied to the data symbols configuring the PLP is larger than that of the FFT applied to the preamble, preamble detection performance can be improved even in a delay spread channel having a length equal to or greater than that of the useful symbol portion A of the preamble. Since both the cyclic prefix (B) and the cyclic suffix (C) are used in the preamble, the fractional carrier frequency offset can be estimated by the correlation process.

FIG. 51 is a view showing an embodiment of detecting a preamble signal shown in FIG. 50 and estimating a timing offset and a frequency offset. This embodiment may be included in the frame detector 221 or the frame synchronization unit 222.

This embodiment may include a first delay unit 601, a complex conjugate calculation unit 603, a first multiplier 605, a second multiplier 607, a first filter 611, a second delay unit 615, a third multiplier 609, a second filter 613, a fourth multiplier 617, a peak search unit 619, and a phase measurement unit 621.

The first delay unit 601 may delay a received signal. For example, the first delay unit 601 may delay the received signal by the length of the useful symbol portion (A) of the first pilot signal.

The complex conjugate calculation unit 603 may calculate the complex conjugate of the delayed first pilot signal and output the calculated signal.

The first multiplier 605 may multiply the signal output from the complex conjugate calculation unit 603 by the received signal and output the multiplied signal.

Since the first pilot signal includes the portions B and C obtained by frequency-shifting the useful portion A, the respective correlation values are obtained by shifting the received signals by the respective frequency shift amounts. In the first pilot signal, the portion B is a portion which is frequency-shifted up or frequency-shifted down from the portion A, and C is a portion which is frequency-shifted up or frequency-shifted down from the portion A.

For example, if the output of the complex conjugate calculation unit 603 is used, the output of the first multiplier 605 may include the correlation result of B (or the complex conjugate of B) and A (or the complex conjugate of A).

The second multiplier 607 may multiply the signal output from the first multiplier 605 by the frequency shift amount (denoted by ej^(fSH)t) applied to the portion B and output the multiplied signal.

The first filter 611 performs a moving average during a predetermined period with respect to the signal output from the second multiplier 607. The moving average portion may be the length of the cyclic prefix (B) or the length of the cyclic suffix (C). In this embodiment, the first filter 611 may calculate an average of the signal included in the length of the portion B. Then, in the result output from the first filter 611, the correlation value of the portions A and C included in the portion, of which the average is calculated, substantially becomes zero and the correlation result of the portions B and A remains. Since the signal of the portion B is multiplied by the frequency shift value by the second multiplier 607, it is equal to the signal obtained by duplicating the second half of the portion A.

The third multiplier 609 may multiply the signal output from the first multiplier 605 by the frequency shift amount (denoted by −ejfSHt) applied to the portion C and output the multiplied signal.

The second filter 613 performs a moving average during a predetermined period with respect to the signal output from the third multiplier 609. The moving average portion may be the length of the cyclic prefix (B) or the length of the cyclic suffix (C). In this embodiment, the second filter 613 may calculate the average of the signal included in the length of the portion C. Then, in the result output from the second filter 613, the correlation value of the portions A and B included in the portion, of which the average is calculated, substantially becomes zero and the correlation result of the portions C and A remains. Since the signal of the portion C is multiplied by the frequency shift value by the third multiplier 609, it is equal to the signal obtained by duplicating the first half of the portion A.

The length T_(B) of the portion of which the moving average is performed by the first filter 611 and the second filter 613 is expressed as follows. T _(B) =k/f _(SH),  [Equation 2]

where, k denotes an integer. In other words, the unit f_(SH) of the frequency shift used in the portions B and C may be decided by k/TB.

The second delay unit 615 may delay the signal output from the first filter 611. For example, the second delay unit 615 delays the signal filtered by the first filter 611 by the length of the portion B and outputs the delayed signal.

The fourth multiplier 617 multiplies the signal delayed by the second delay unit 615 by the signal filtered by the second filter 613 and outputs the multiplied signal.

The peak search unit 619 searches for the location where a peak value is generated from the multiplied signal output from the fourth multiplier 617 and outputs the searched location to the phase measurement unit 621. The peak value and the location may be used for the timing offset estimation.

The phase measurement unit 621 may measure the changed phase using the peak value and the location output from the peak search unit 619 and output the measured phase. The phase value may be used for the fractional carrier frequency offset estimation.

Meanwhile, an oscillator for generating the frequency used for performing the frequency shift by the second multiplier 607 and the third multiplier 609 may generate any phase error.

Even in this case, the fourth multiplier 617 can eliminate the phase error of the oscillator. The results output from the first filter 611 and the second filter 613 and the result output from the fourth multiplier 617 may be expressed by the following equation. y _(MAF1) =∥a ₁(n)∥² ·e ^(j2πΔ) ^(f) ^(+θ)  [Equation 3] y _(MAF2) =∥a ₂(n)∥² ·e ^(j2πΔ) ^(f) ^(−θ) y _(prod) =∥a ₁(n)∥² ·∥a ₂(n)∥² ·e ^(j2π·2Δ) ^(f)

where, y_(MAF1) and y_(MAF2) respectively denote the outputs of the first filter 611 and the second filter 613, and yProd denotes the output of the fourth multiplier 617. In addition, a1 and a2 respectively denote the levels of the correlation results and Δf and θ respectively denote the frequency offset and the phase error of the oscillator.

Accordingly, y_(MAF1) and y_(MAF2) may include the phase errors of the oscillator having different signs, but the phase error of the oscillator is eliminated in the result of the fourth multiplier 617. Accordingly, the frequency offset Af can be estimated regardless of the phase error of the oscillator of the signal receiving apparatus.

The estimated frequency offset may be expressed by the following equation. f _(B) =∠y _(prod)/4π  [Equation 4]

where, the estimated frequency offset Δf is 0<=Δf<0.5.

FIG. 52 is a view showing another embodiment of the structure of the first pilot signal. In the first pilot signal, the frequency shift of the first half of the useful portion A is the cyclic prefix (B) and the frequency shift of the second half of the useful portion A is the cyclic suffix (C). The lengths of the useful portion A for generating the portions B and C may be, for example, ½ of the length of the portion A, and the lengths of B and C may be different.

FIG. 53 is a view showing an embodiment of detecting the first pilot signal shown in FIG. 52 and measuring a timing offset and a frequency offset using the detected result. In this embodiment, for convenience of description, B and C respectively denote the cyclic prefix and the cyclic suffix obtained by frequency-shifting ½ of the length of the portion A.

This embodiment includes a first delay unit 601, a complex conjugate calculation unit 603, a first multiplier 605, a second multiplier 607, a first filter 611, a second delay unit 615, a third multiplier 609, a second filter 613, a fourth multiplier 617, a peak search unit 619, and a phase measurement unit 621. That is, this embodiment is equal to the embodiment of FIG. 51, but the features of the components may be changed according to the length of the portion A by which the portions B and C are generated. B denotes a portion frequency-shifted down from the portion A, and C denotes a portion frequency-shifted up from the portion A.

The first delay unit 601 may delay a received signal. For example, the first delay unit 601 may delay the received signal by ½ of the length of the useful symbol portion A of the first pilot signal.

The complex conjugate calculation unit 603 may calculate the complex conjugate of the delayed first pilot signal and output the calculated signal.

The first multiplier 605 may multiply the signal output from the complex conjugate calculation unit 603 by the received signal and output the multiplied signal.

The second multiplier 607 may multiply the signal output from the first multiplier 605 by the frequency shift amount (denoted by ejfSHt) applied to the portion B and output the multiplied signal.

The first filter 611 performs a moving average during a predetermined period with respect to the signal output from the second multiplier 607. The moving average portion may be the length of the cyclic prefix (B). In this embodiment, the first filter 611 may calculate the average of the signal included in the length of the portion B. Then, in the result output from the first filter 611, the correlation value of the portions A and C included in the portion, of which the average is calculated, substantially becomes zero and the correlation result of the portions B and A remains. Since the signal of the portion B is multiplied by the frequency shift value by the second multiplier 607, it is equal to the signal obtained by duplicating the second half of the portion A.

The third multiplier 609 may multiply the signal output from the first multiplier 605 by the frequency shift amount (denoted by −ejfSHt) applied to the portion C and output the multiplied signal.

The second filter 613 performs a moving average during a predetermined period with respect to the signal output from the third multiplier 609. The moving average portion may be the length of the cyclic suffix (C). In this embodiment, the second filter 613 may calculate the average of the signal included in the length of the portion C. Then, in the result output from the second filter 613, the correlation value of A and B included in the portion, of which the average is calculated, substantially becomes zero and the correlation result of the portions C and A remains. Since the signal of the portion C is multiplied by the frequency shift value by the third multiplier 609, it is equal to the signal obtained by duplicating the first half of the portion A.

The second delay unit 615 may delay the signal output from the first filter 611. For example, the second delay unit 615 delays the signal filtered by the first filter 611 by the length of the portion B+½A and outputs the delayed signal.

The fourth multiplier 617 multiplies the signal delayed by the second delay unit 615 by the signal filtered by the second filter 613 and outputs the multiplied signal.

The peak search unit 619 searches for the location where a peak value is generated from the multiplied signal output from the fourth multiplier 617 and outputs the searched location to the phase measurement unit 621. The peak value and the location may be used for the timing offset estimation.

The phase measurement unit 621 may measure the changed phase using the peak value and the location output from the peak search unit 619 and output the measured phase. The phase value may be used for the fractional carrier frequency offset estimation.

As described above, an oscillator for generating the frequency used for performing the frequency shift by the second multiplier 607 and the third multiplier 609 may generate any phase error. However, even in this embodiment, the fourth multiplier 617 can eliminate the phase error of the oscillator.

The results output from the first filter 611 and the second filter 613 and the result output from the fourth multiplier 617 may be expressed by the following equation. y _(MAF1) =∥a ₁(n)∥² ·e ^(j2πΔ) ^(f) ^(+θ)  [Equation 5] y _(MAF2) =∥a ₂(n)∥² ·e ^(j2πΔ) ^(f) ^(−θ) y _(prod) =∥a ₁(n)∥² ·∥a ₂(n)∥² ·e ^(j2π·2Δ) ^(f)

where, y_(MAF1) and y_(MAF2) respectively denote the outputs of the first filter 611 and the second filter 613, and yProd denotes the output of the fourth multiplier 617. In addition, a1 and a2 respectively denote the levels of the correlation results and Δf and θ respectively denote the frequency offset and the phase error of the oscillator.

Accordingly, y_(MAF1) and y_(MAF2) may include the phase errors of the oscillator having different signs, but the phase error of the oscillator is eliminated in the result of the fourth multiplier 617. Accordingly, the frequency offset Af can be estimated regardless of the phase error of the oscillator of the signal receiving apparatus.

The estimated frequency offset may be expressed by the following equation. f _(B) =∠y _(prod)/2π  [Equation 6]

where, the estimated frequency offset Δf is 0<=Δf<1.

That is, phase aliasing may be generated in a range of 0.5<=Δf<1 in the frequency offset estimated in [Equation 4], but phase aliasing is not generated in the frequency offset estimated in [Equation 6]. Accordingly, the frequency offset can be more accurately measured. The structure of the first pilot signal may be used in the data symbol and the second frequency signal. If such a structure is used, offset estimation performance such as CW interference can be improved and the reception performance of the receiver can be improved.

FIG. 54 is a view showing an embodiment of detecting the first pilot signal and measuring a timing offset and a frequency offset using the detected result.

This embodiment includes a first delay unit 601, a third delay unit 602, a first complex conjugate calculation unit 603, a second complex conjugate calculation unit 604, a first multiplier 605, a fifth multiplier 606, a second multiplier 607, a first filter 611, a second delay unit 615, a third multiplier 609, a second filter 613, a fourth multiplier 617, a peak search unit 619, and a phase measurement unit 621.

In this embodiment, the first delay unit 601 may delay a received signal. For example, the first delay unit 601 may delay the received signal by the length of the cyclic suffix.

The third delay unit 602 may delay the signal delayed by the first delay unit 601. For example, the third delay unit 602 further delays the signal by a difference between the length of the cyclic prefix and the length of the cyclic suffix.

The first complex conjugate calculation unit 603 may calculate the complex conjugate of the signal delayed by the third delay unit 602 and output the calculated signal. The second complex conjugate calculation unit 604 may calculate the complex conjugate of the signal delayed by the first delay unit 601 and output the calculated signal.

The first multiplier 605 may multiply the signal output from the first complex conjugate calculation unit 603 by the received signal and output the multiplied signal. The fifth multiplier 606 may multiply the complex conjugate calculated by the second complex conjugate calculation unit 604 by the received signal and output the multiplied signal.

The second multiplier 607 may multiply the signal output from the first multiplier 605 by the frequency shift amount (denoted by e^(jfSHt)) applied to the portion B and output the multiplied signal.

The first filter 611 performs a moving average during a predetermined period with respect to the signal output from the second multiplier 607. The moving average portion may be the length of the useful portion (A) of the first pilot signal.

The third multiplier 609 may multiply the signal output from the second multiplier 604 by the frequency shift amount (denoted by −e^(jfSHt)) applied to the portion C and output the multiplied signal.

The second filter 613 performs a moving average during a predetermined period with respect to the signal output from the third multiplier 609. The moving average portion may be the length of the useful portion A of the first pilot signal.

The second delay unit 615 may delay the signal output from the first filter 611. For example, the second delay unit 615 delays the signal filtered by the first filter 611 by the length of the useful portion (A) of the first pilot signal and outputs the delayed signal.

The fourth multiplier 617 multiplies the signal delayed by the second delay unit 615 by the signal filtered by the second filter 613 and outputs the multiplied signal. The fourth multiplier 617 may eliminate the phase error of the oscillator.

The operations of the peak search unit 619 and the phase measurement unit 621 are equal to those of the above-described embodiment. The peak search unit 619 searches for the location where a peak value is generated from the multiplied signal output from the fourth multiplier 617 and outputs the searched location to the phase measurement unit 621. The peak value and the location may be used for the timing offset estimation.

FIG. 55 is a view showing an embodiment of a method of transmitting a signal.

A service stream is converted to a PLP (S110). The PLP can be generated by modulating a service stream such as a transport stream and a GSE packet, in which error correction encoding and symbol mapping are performed on the service stream. The modulated service stream can be distributed in at least one signal frame and can be transmitted over at least one physical channel as a PLP. For example, a process of modulating a service stream to a PLP can be performed by following steps S110 a to S110 d.

A service stream such as a transport stream and a GSE packet transferring service is error-correction-coded (S110 a). An error correction coding scheme may be changed according to the service streams.

An LDPC error correction coding scheme may be used as the error correction coding scheme and the error correction coding may be performed at various code rates. The bits which are error-correction-coded according to a specific error correction code rate may be included in an error correction coded block according to the error correction coding mode. If the error correction coding scheme is the LDPC, a normal mode (64800 bits) and a short mode (16200 bits) may be used.

The error-correction-coded service stream is interleaved (S110 b). The interleaving may be performed by differentiating the directions for writing and reading the bits included in the error correction coded block in and from a memory. The number of rows and the number of columns of the memory may be changed according to the error correction coding mode. The interleaving may be performed in the unit of the error correction coded blocks.

The interleaved bits of the service stream are mapped to symbols (S110 c). A symbol mapping method may be changed according to service streams or in the service stream. For example, as the symbol mapping method, a higher order symbol mapping method and a lower order symbol mapping method may be used. When the symbols are mapped, the interleaved bits of the service stream may be demultiplexed according to the symbol mapping method or the code rate of the error correction code, and the symbols may be mapped using the bits included in the demultiplexed sub streams. Then, the sequence of the bits in the cell word mapped to the symbols may be changed.

The mapped symbols are interleaved (S110 d). The mapped symbols may be interleaved in the unit of error correction coded blocks. Time interleavers 132 a and 132 b may interleave the symbols in the unit of error correction coded blocks. That is, the service stream is interleaved again in the symbol level.

The PLP converted as described above is allocated in at least one signal frame and a preamble including a first pilot signal is arranged in a beginning part of the signal frame (S150). The allocation of the PLP may be described as follow.

The interleaved symbols of the service stream are split, the split symbols are allocated to a signal frame having at least one frequency band and including slots which are temporally split in the frequency bands, and a preamble including a first pilot signal is arranged in a start portion of the signal frame. The interleaved symbols of the service stream may configure the PLP with respect to the service stream for providing the service. The symbols configuring the PLP may be split and allocated to the signal frame. The PLP may be allocated to at least one signal frame having at least one frequency band. If a plurality of frequency bands is arranged, the symbols configuring the PLP may be arranged in the slots shifted between the frequency bands. The bits included in the service stream may be arranged in the signal frame in the unit of interleaved error correction coded blocks.

The signal frame is converted into a time domain according to an OFDM scheme (S160).

The cyclic prefix obtained by frequency-shifting a first portion of an useful portion of the first pilot signal and the cyclic suffix obtained by frequency-shifting a second portion of the useful portion are inserted into the first pilot signal in the time domain (S170). If the preamble is not inserted in the frequency domain, the preamble including the first pilot signal and the second pilot signal may be inserted in the time domain. The first pilot signal of the time domain may include the useful portion, the cyclic prefix of the first portion of the useful portion and the cyclic suffix of the second portion of the useful portion. The first portion may be a backmost portion or the foremost portion of the useful portion. The second portion may be the foremost portion or the backmost portion of the useful portion.

The signal frame including the first frame signal is transmitted over at least one RF channel (S180).

Since the useful portion of the first pilot signal includes the frequency-shifted cyclic prefix and cyclic suffix, the signal frame can be clearly identified as the structure of the first pilot signal. The timing offset or the frequency offset may be estimated and compensated for using the structure of the first pilot signal.

FIG. 56 is a view showing an embodiment of a method of receiving a signal.

A signal is received from a specific frequency band transferring signal frames (S210). The signal frame may be transmitted over at least one frequency band. The signal may be received from a specific frequency band

From the received signal, a first pilot signal including a cyclic prefix obtained by frequency-shifting a first portion of an useful portion and a cyclic suffix obtained by frequency-shifting a second portion of the useful portion is identified, and the signal frame including the PLPs is identified is demodulated by the OFDM scheme using the first pilot signal (S220). The demodulating process using information set in the first pilot signal will be described in detail later.

The identified signal frame is parsed (S230). The signal frame may include at least one frequency band. In the signal frame, a first PLP including the error correction coded blocks of the symbols, to which the service stream is mapped, may be allocated to OFDM symbols together with a second PLP including the error correction coded blocks of another service stream. If the signal frame includes a plurality of frequency bands, the error correction coded blocks of the PLP may be allocated to the OFDM symbols which are temporally shifted in the plurality of frequency bands.

A service can be obtained from the PLP of the parsed signal frame (S240), in which this process is described in steps S240 a to S240 c.

The symbols, to which the service stream is mapped, are deinterleaved from the parsed signal frame (S240 a). The deinterleaving may be performed in the symbol level which the service stream is mapped to. For example, the time deinterleavers 245 a and 245 b may deinterleave the error correction coded blocks including the symbols, to which the service stream is mapped.

Then, the deinterleaved symbols are demapped so as to obtain the service stream (S240 b). When the symbols are demapped, a plurality of sub streams obtained by demapping the symbols may be output, the output sub streams may be multiplexed, and the error-correction-coded service stream may be output. The multiplexing scheme may be changed according to the symbol mapping method and the error correction code rate. The symbol demapping method may be changed in one service stream or according to service streams.

The service stream is deinterleaved and the deinterleaved service stream is error-correction-coded (240 c).

According to an apparatus for transmitting and receiving a signal and a method for transmitting and receiving a signal of an embodiment of the present invention, it is possible to readily detect and restore a transmitted signal. In addition, it is possible to improve the signal transmission/reception performance of the transmitting/receiving system.

FIG. 57 is a flowchart illustrating an embodiment of identifying a first pilot signal and estimating an offset in a demodulating process.

The first pilot signal includes the cyclic prefix obtained by frequency-shifting the first portion of the useful portion thereof and the cyclic suffix obtained by frequency-shifting the second portion of the useful portion thereof. The timing offset and the frequency offset may be calculated using the first pilot signal as follows.

The received signal is delayed (S311). For example, the delay portion may be the useful portion of the first pilot signal or ½ of the useful portion. Alternatively, the delay portion may be the length of the cyclic prefix or the length of the cyclic suffix.

The complex conjugate of the delayed signal is calculated (S313).

The complex conjugate of the received signal and the delayed signal are multiplied (S315). The delayed signal multiplied by the complex conjugate may be the signal having the above-described length. If the delay signal is the length of the cyclic prefix or the cyclic suffix, the complex conjugate of the delayed signal may be calculated.

The signal multiplied by the complex conjugate is inversely shifted according to the frequency shift of the cyclic prefix (S317). That is, the signal multiplied by the complex conjugate is shifted by the inverse shift amount of the frequency shift amount of the cyclic prefix signal. That is, a signal which is frequency shifted up is frequency shifted down (or the signal which is frequency shifted down is frequency shifted up).

Then, an average is calculated with respect to the signal which is inversely shifted according to the frequency shift of the cyclic prefix (S319). The portion of which the average is calculated may be the length of the cyclic prefix or the length of the useful portion A of the first pilot signal depending on the embodiments. Since the average is calculated with respect to the signal having the same length along with the received signal, the moving average value may be output along with the received signal.

The signal of which the average is calculated is delayed (S321). The delay portion may be the sum of the length of the cyclic prefix and the length of ½ of the useful period, the length of the cyclic prefix, or the length of the useful portion A of the first pilot signal, according to the embodiment.

The signal multiplied in the step S315 is inversely shifted according to the frequency shift of the cyclic suffix (S323). The signal multiplied by the complex conjugate is shifted by the inverse shift amount of the frequency shift amount of the cyclic suffix signal. That is, a signal which is frequency shifted up is frequency shifted down (or the signal which is frequency shifted down is frequency shifted up).

An average is calculated with respect to the signal which is inversely shifted according to the frequency shift of the cyclic suffix (S325). The moving average is performed with respect to the signal corresponding to the length of the calculated cyclic suffix or the length of the useful portion of the first pilot signal according to the embodiments.

The signal delayed in the step S321 and the signal of which the average is calculated in the step S325 are multiplied (S327).

A peak location of the multiplied result is searched for (S329) and the phase of the signal is measured using the peak (S331). The searched peak may be used for estimating the timing offset and the measured phase may be used for estimating the frequency offset.

In this flowchart, the length of the cyclic suffix, the length of the cyclic prefix and the frequency inverse shift amount may be changed.

According to the apparatus for transmitting and receiving the signal and the method for transmitting and receiving the signal of the invention, if the data symbol configuring the PLP and the symbols configuring the preamble are modulated in the same FFT mode, the probability that the data symbol is detected by the preamble is low and the probability that the preamble is erroneously detected is reduced. If continuous wave (CW) interference is included like the analog TV signal, the probability that the preamble is erroneously detected by a noise DC component generated at the time of correlation is reduced.

According to the apparatus for transmitting and receiving the signal and the method for transmitting and receiving the signal of the invention, if the size of the FFT applied to the data symbol configuring the PLP is larger than that of the FFT applied to the preamble, the preamble detecting performance may be improved even in a delay spread channel having a length equal to or greater than that of the useful symbol portion A of the preamble. Since both the cyclic prefix (B) and the cyclic suffix (C) are used in the preamble, the fractional carrier frequency offset can be estimated.

The disclosed structure of the pilot signal may not be used for a signal frame including the PLP, and if the pilot signal is used for any signal frame, the described effect can be taken.

MODE FOR THE INVENTION

The embodiments of the invention are described in the best mode of the invention.

INDUSTRIAL APPLICABILITY

A method of transmitting/receiving a signal and an apparatus for transmitting/receiving a signal of the present invention can be used in broadcast and communication fields. 

What is claimed is:
 1. A method of transmitting a broadcast signal, the method comprising: building a signal frame based on service data; modulating the signal frame according to an orthogonal frequency division multiplexing (OFDM) scheme; inserting a symbol into a beginning part of the modulated signal frame; and transmitting the broadcast signal including the signal frame and the inserted symbol, wherein the symbol comprises an effective portion, a cyclic prefix obtained by frequency-shifting a first portion of the effective portion, and a cyclic suffix obtained by frequency-shifting a second portion of the effective portion, wherein the first portion is a foremost part of the effective portion and the second portion is a backmost part of the effective portion.
 2. An apparatus for transmitting a broadcast signal, the apparatus comprising: means for building a signal frame based on service data; means for modulating the signal frame according to an orthogonal frequency division multiplexing (OFDM) scheme; means for inserting a symbol into a beginning part of the modulated signal frame; and means for transmitting the broadcast signal including the signal frame and the inserted symbol, wherein the symbol comprises an effective portion, a cyclic prefix obtained by frequency-shifting a first portion of the effective portion, and a cyclic suffix obtained by frequency-shifting a second portion of the effective portion, wherein the first portion is a foremost part of the effective portion and the second portion is a backmost part of the effective portion. 